System with phase jumping locked loop circuit

ABSTRACT

An integrated circuit device having a select circuit, a summing circuit and a phase mixer. The select circuit selects one of a plurality of offset values as a selected offset. The summing circuit sums the selected offset with a phase count value, the phase count value indicating a phase difference between a reference clock signal and a first plurality of clock signals. The phase mixer combines the first plurality of clock signals in accordance with the sum of the selected offset and the phase count value to generate an output clock signal.

CROSS-REFERENCE TO RELATED APPLICATIONS

(P117) This application is a continuation-in-part of U.S. patentapplication Ser. No. 10/104,230, filed Mar. 22, 2002, and claims thebenefit of U.S. Provisional Application No. 60/408,063, filed Sep. 3,2002, U.S. Provisional Application No. 60/408,101 filed Sep. 3, 2002,and U.S. Provisional Application No. 60/436,745 filed Dec. 27, 2002.Each of U.S. Provisional Application Nos. 60/408,063, 60/408,101 and60/436,745 is hereby incorporated by reference in its entirety.

FIELD OF THE INVENTION

The present invention relates generally to the field of high-speedsignaling, and more particularly to timing signal generation within adelay-locked loop or phase-locked loop circuit.

BACKGROUND

Delay-locked loop (DLL) circuits are often used in high-speed signalingsystems to generate signals for precisely timing sampling andtransmission events within input/output circuits. FIG. 1 illustrates aprior-art delay-locked loop (DLL) circuit 100 that includes a referenceloop 101, tracking loop 103 and clock generator 105. A complementarypair of reference clock signals, CLK and /CLK (102 and 104), aresupplied to the reference loop 101 which, in turn, generates eightincrementally delayed clock signals 122, t₀-t₃ and /t₀-/t₃, referred toas phase vectors. Ideally the phase vectors are evenly phase-spacedwithin a time interval that corresponds to a cycle of the referenceclock signal 102 such that a 45° phase offset separates eachphase-adjacent pair of phase vectors. The tracking loop 123 includes amixer 117, clock tree circuit 119 and phase detector 115 which cooperateto generate a feedback clock signal 112 that is phase aligned with thereference clock signal 102. The mixer 117 receives the phase vectors 122from the reference loop 101 and interpolates between a selected pair ofthe phase vectors to generate a mix clock signal 110. The mix clocksignal 110 propagates through the clock tree circuit 119 (typically aset of amplifiers used to generate multiple instances of the mix clocksignal 110) to generate the feedback clock signal 112. The phasedetector 115 compares the feedback clock signal 112 with the referenceclock signal 102 and generates a phase adjust signal 106 (U/D) accordingto which clock signal leads the other. For example, if the referenceclock 102 signal leads the feedback clock signal 112, the phase detector115 signals the mixer 117 (i.e., by appropriate state of the phaseadjust signal) to shift interpolation toward the leading one of theselected phase vectors and away from the trailing phase vector, therebyadvancing the phase of the feedback clock 112 and reducing the phasedifference between the reference and feedback clock signals. If thereference clock signal 112 still leads (or lags) the feedback clocksignal after interpolation has been shifted completely to one of theselected phase vectors, a different pair of phase vectors (i.e.,bounding an adjacent phase range) is selected by the mixer 117. The DLLcircuit 100 achieves phase lock when the phase of the feedback clocksignal 112 becomes aligned with the phase of the reference clock signal102.

The clock generator 105 includes a mixer 121 and clock tree circuit 123that mirror the operation of the mixer 117 and clock tree circuit 119within the tracking loop 103 to generate a local clock signal 116(LCLK). The mixer 121 receives the phase adjust signal 106 generatedwithin the mix loop 103 and therefore, when an offset control value 108(OFFSET) is zero, performs nominally the same interpolation operation onthe same pair of selected vectors as the mixer 117. Ideally, as theadjust signal 106 is incremented and decremented, the mixer 121 tracksthe operation of the mixer 117 such that the local clock signal 116 andthe feedback 112 are phase aligned. The offset control value 106 issummed with a count value maintained within the mixer 121 to provide acontrolled, adjustable offset between the local clock signal 116 andreference clock signal 112, thereby allowing compensation for skewbetween the reference clock signal and a sampling instant, transmitinstant or other event to be timed by the local clock signal 116.

FIG. 2 illustrates a prior-art phase mixer 121 in greater detail. Themixer 121 includes a counter 139, adder 141, bias voltage generator 143,and a bank of differential amplifiers 151. Each of the differentialamplifiers 151 is formed by a pair of differentially coupled transistorshaving gate terminals coupled to receive a respective pair ofcomplementary phase vectors, source terminals coupled to the drainterminal of a corresponding biasing transistor 153, and drain terminalscoupled to a mix clock line 116 and complement mix clock line 118,respectively. The mix clock line 116 and complement mix clock lines arepulled up to a supply voltage via respective resistive elements, R. Bythis arrangement, when a given one of the biasing transistors 153 isbiased to a current conducting state, the corresponding differentialamplifier is enabled to draw current via resistive elements R inaccordance with the input phase vectors, thereby causing the phasevector and its complement to appear on the complement mix clock line 118and mix clock line 116 as a mix clock signal (MCLK) and complement mixclock signal (/MCLK), respectively. When two of the biasing transistors153 are biased to a current conducting state, the input phase vectorssupplied to the corresponding differential amplifiers are each enabledto contribute to the mix clock signal. The mix clock signal willinitially slew (i.e., transition between states) at a rate determined bya leading one of the input phase vectors and then, after the trailingvector begins to transition, at a rate determined by the sum of theleading and trailing phase vectors, thereby yielding a mix clock signalphase that lies between the leading and trailing vectors according tothe relative bias currents drawn by the biasing transistors 153.

The counter 171 is incremented and decremented in response to the phaseadjust signal 106, and summed with the offset value 108 in adder circuit141 to generate a phase control word 142. The phase control word 142 isdecoded by decode logic 145 within the bias voltage generator 143 togenerate a complementary pair of bias words 146 which are supplied to adigital-to-analog converter (DAC) 147. The most significant three bitsof the complementary control values 146 indicate one of eightphase-adjacent pairs of phase vectors to be mixed to generate the mixclock signal, MCLK, and corresponding complementary phase vectors to bemixed to generate the complementary mix clock signal, /MCLK. Thus, theDAC 147 generates bias voltages on bias lines 154 in response to thecomplementary control values 146, such that at most two of the biasingtransistors 153 are enabled at any given time, all other biasingtransistors 153 being placed in a non-conducting state. As the countvalue is incremented by the counter, the bias voltage applied to one ofthe two enabled biasing transistors is increased, increasing thecontribution of the corresponding phase vector to phase of the mix clocksignal, and the bias voltage applied to the other selected biasingtransistor is decreased, decreasing the contribution of thecorresponding phase vector to the mix clock signal. Thus, as the countvalue is incremented and decremented, the phase of the mix clock signalis correspondingly advanced and delayed.

Because of the relatively small voltage steps generated by the DAC 147and the high impedance load presented by the gates of biasingtransistors 153, substantial time is typically required for eachstepwise change in the output of DAC 147 to settle and produce a stablemix clock signal. Also, noise on the bias voltage lines 154 tends toproduce phase jitter in the mix clock signals 116 and 118 so thatcapacitive elements are typically coupled to the bias voltage lines 154as illustrated by (i.e., as illustrated by capacitive element, C, inFIG. 2). Unfortunately, capacitive loading of the bias voltage lines 154further increases the time required for the lines 154 to settle inresponse to an increase or decrease of the bias voltage. Additionally,significant changes in the RC time constant result from processvariations and from changes in temperature and voltage, making itdifficult to quantify or predict the worst case settling time for thebias voltage lines 154. Consequently, several cycles of the referenceclock signal are typically required for the phase of the output clocksignal to stabilize in response to each bias voltage change. This is asignificant disadvantage of the mixer 121, as a relatively long time istypically required to perform a phase locking operation in whichnumerous successive phase steps are needed to reach phase lock. Theability to rapidly switch between phase offsets in response to changesin the offset control value 108 is similarly limited by the DAC settlingtime.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention is illustrated by way of example, and not by wayof limitation, in the figures of the accompanying drawings and in whichlike reference numerals refer to similar elements and in which:

FIG. 1 illustrates a prior art delay-locked loop circuit;

FIG. 2 illustrates a prior-art phase mixer;

FIG. 3 illustrates a phase-jumping locked loop circuit according to anembodiment of the invention;

FIG. 4 illustrates a phase-jumping mixer according to an embodiment ofthe invention;

FIG. 5 illustrates the correspondence between range select values, phasevectors and mix-enabled differential amplifiers in the phase-jumpingmixer of FIG. 4;

FIG. 6 illustrates the even- and odd-phase current DACs of FIG. 4according to an embodiment of the invention;

FIG. 7 illustrates an alternative amplifier biasing circuit according toan alternative embodiment of the invention;

FIG. 8 illustrates the phase steps produced in a mix clock signal as theinterpolation weight generated by the mix logic of FIG. 4 is incrementedfrom zero to a maximum value;

FIG. 9 illustrates a mixer according to an embodiment of the inventionin which the resistance values of mix clock line pull-up elements aredynamically adjusted to maintain a relatively constant mix clock signalswing over changes in bias current drawn by an amplifier biasingcircuit;

FIG. 10 illustrates a reference loop according to an embodiment of theinvention for outputting process-, temperature- and voltage-trackingbias voltages to the phase-jumping mixers of FIGS. 4 and 9;

FIG. 11 illustrates an exemplary embodiment of a delay element that maybe used within the reference loop of FIG. 10;

FIG. 12 illustrates the voltage generation circuit of FIG. 10 accordingto an embodiment of the invention;

FIG. 13 illustrates the dual-control current source of FIG. 12 accordingto an embodiment of the invention;

FIG. 14 illustrates the manner in which the dual-control current sourceof FIG. 12 may be used to achieve a desired operation over processvariations that range between fast and slow corners;

FIG. 15 illustrates a system in which a phase jumping locked loopcircuit according to embodiments described above in reference to FIGS.3-14 may be used;

FIG. 16 illustrates a signaling device according to an embodiment of theinvention; FIG. 17 illustrates the relationship between an exemplarydata waveform, the direction signal of FIG. 16 and correspondingtransitions of the transmit and receive clock signals generated on theshared clock line of FIG. 16;

FIG. 18 illustrates an alternative circuit arrangement for generating aphase control value within the offset clock generator of FIG. 16;

FIG. 19 illustrates a binary phase searching operation according to anembodiment of the invention;

FIG. 20 illustrates possible phase relationships between a referenceclock signal and a feedback clock signal generated by a tracking loop;

FIG. 21 illustrates a tracking loop for performing a phase searchingoperation according to an embodiment of the invention;

FIG. 22 is a flow diagram of a phase searching operation illustratedaccording to an embodiment of the invention;

FIG. 23 illustrates the phase offset of an incoming data eye relative tothe feedback clock signal generated by a tracking loop;

FIG. 24 illustrates a division of a cycle of a feedback clock signalinto exemplary search ranges;

FIG. 25 is a flow diagram of a coarse linear search for leading andtrailing edges of a data eye according to an embodiment of theinvention;

FIG. 26 illustrates a binary search for an edge of a data eye accordingto an embodiment of the invention;

FIG. 27 illustrates a timing maintenance operation according to anembodiment of the invention;

FIG. 28 illustrates a circuit for generating a phase control valueaccording to an embodiment of the invention;

FIG. 29 is a flow diagram of a timing maintenance operation according toan embodiment of the invention;

FIG. 30 illustrates a signaling device according to an embodiment of theinvention;

FIG. 31 illustrates another phase-jumping locked loop circuit accordingto an embodiment of the invention;

FIG. 32 illustrates an exemplary relationship between the offset select,mix clock, and hold clock signals of FIG. 31;

FIG. 33 illustrates the clock hold circuit of FIG. 31 according to anembodiment of the invention;

FIG. 34 is an exemplary state diagram of the hold control circuit ofFIG. 33;

FIG. 35 illustrates exemplary timing relationships between a clockwindow and a jump window;

FIG. 36 is an exemplary state diagram of the keepout circuit of FIG. 33;and FIG. 37 illustrates an exemplary embodiment of the synchronizinglogic of FIG. 33.

DETAILED DESCRIPTION

In the following description and in the accompanying drawings, specificterminology and drawing symbols are set forth to provide a thoroughunderstanding of the present invention. In some instances, theterminology and symbols may imply specific details that are not requiredto practice the invention. For example, the interconnection betweencircuit elements or circuit blocks may be shown or described asmulti-conductor or single conductor signal lines. Each of themulti-conductor signal lines may alternatively be single-conductorsignal lines, and each of the single-conductor signal lines mayalternatively be multi-conductor signal lines. Signals and signalingpaths shown or described as being single-ended may also be differential,and vice-versa. Similarly, signals described or depicted as havingactive-high or active-low logic levels may have opposite logic levels inalternative embodiments. As another example, circuits described ordepicted as including metal oxide semiconductor (MOS) transistors mayalternatively be implemented using bipolar technology or any othertechnology in which a signal-controlled current flow may be achieved.With respect to terminology, a signal is said to be “asserted” when thesignal is driven to a low or high logic state (or charged to a highlogic state or discharged to a low logic state) to indicate a particularcondition. Conversely, a signal is said to be “deasserted” to indicatethat the signal is driven (or charged or discharged) to a state otherthan the asserted state (including a high or low logic state, or thefloating state that may occur when the signal driving circuit istransitioned to a high impedance condition, such as an open drain oropen collector condition). A signal driving circuit is said to “output”a signal to a signal receiving circuit when the signal driving circuitasserts (or deasserts, if explicitly stated or indicated by context) thesignal on a signal line coupled between the signal driving and signalreceiving circuits. A signal line is said to be “activated” when asignal is asserted on the signal line, and “deactivated” when the signalis deasserted. Additionally, the prefix symbol “/” attached to signalnames indicates that the signal is an active low signal (i.e., theasserted state is a logic low state). A line over a signal name (e.g.,‘{overscore (<signal name>)}’) is also used to indicate an active lowsignal.

Phase Jumping Locked Loop

A locked loop circuit that enables rapid output clock phase changes,referred to herein as “phase jumping,” is disclosed in variousembodiments. In one embodiment, the locked loop circuit includes one ormore phase-jumping mixers to enable rapid mixing of selected phasevector pairs. Each phase jumping mixer includes a bank ofswitch-selectable differential amplifiers coupled in series with adigitally-controlled current source. The digitally-controlled currentsource may be rapidly switched between different biasing levels toachieve rapid phase transitions (i.e., phase jumps) in a resultant, mixclock signal. In one embodiment, synchronizing elements are provided toensure that digital control signals used to bias thedigitally-controlled current source transition in synchronism (orsubstantially in synchronism), thereby reducing switching noise thatotherwise may produce jitter in the mix clock signal. Also, in oneembodiment, process-, voltage- and temperature-tracking control signalsare generated within a reference loop of the locked loop circuit andused to maintain a desired slew rate and/or amplitude of the signalsthat are combined to generate the mix clock signal, thereby maintainingphase step linearity despite process variations and changes in voltageand temperature.

Overview of a Locked Loop Circuit According to an Embodiment of theInvention

FIG. 3 illustrates a phase-jumping locked loop (PJLL) circuit 200according to an embodiment of the invention. The locked loop circuit 200includes a reference loop 201, tracking loop 203 and offset clockgenerator 205. Complementary reference clock signals 202 and 204 areinput to the reference loop where they propagate through a series ofdelay elements to produce a set of substantially evenly spaced phasevectors 222 (note that even spacing between phase vectors is notrequired; the phase vectors may be unevenly spaced in alternativeembodiments). The reference loop 201 outputs the phase vectors 222 torespective phase jumping mixers 217 and 221 within the tracking loop 203and offset clock generator 205. In the embodiment of FIG. 3, thereference loop generates eight phase vectors, spaced evenly at 45° phaseintervals over a cycle time of the reference clock signal. Inalternative embodiments, more or fewer phase vectors may be generated bythe reference loop 201 (and used in downstream circuits such as mixers217 and 221) such that the phase range between adjacent phase vectors isless than or greater than 45°. In one embodiment, the reference loop 201outputs a slew control signal 226 and amplitude control signal 228 tothe phase-jumping mixers 217 and 221 to maintain substantially linearphase mixing and output clock amplitude through changes in voltage andtemperature, and over fast and slow process comers.

The tracking loop 203 includes a clock tree circuit 219, phase detector247 and phase counter 225, along with the phase-jumping mixer 217. A mixclock signal 210 generated by the phase-jumping mixer propagates throughthe clock tree circuit 219 to generate a feedback clock signal 212 whichis provided, in turn, to the phase detector 247. The phase detector 247compares the feedback clock signal to the reference clock signal 202 andoutputs a phase adjust signal 206 to the phase counter 225 according towhich of the clock signals 202 and 204 leads the other. The phasecounter 225 increments and decrements a phase count value 230 inresponse to the phase adjust signal 206. The phase count value 230represents a phase offset between a selected one of the phase vectors222 (e.g., one of the phase vectors designated to be a 0° vector) andthe reference clock signal 202, and is supplied to the phase jumpingmixer 217 and to the offset clock generator 205. The phase-jumping mixer217 selects and interpolates between a pair of phase vectors accordingto the phase count value, thereby advancing or retarding the phase ofthe mix clock signal 210 in response to decreases and increases in thephase count value 230. Thus, the phase detector 247, phase counter 225,and phase jumping mixer 217 form a closed-loop, negative-feedbackcircuit that adjusts the phase of the feedback clock signal 212 asnecessary to reduce the phase difference between the feedback clocksignal 212 and the reference clock signal 202.

The offset clock generator includes a phase jumping mixer 221, clocktree 223 and adder 235. The adder 235 generates a phase control word 232by summing the phase count value 230 from the tracking loop 203 with anoffset control value 208 (OFFSET). In one embodiment, the offset controlvalue 208 is supplied by other logic within the integrated circuit thatcontains the locked loop circuit 200 (e.g., a configuration register orbank of configuration registers). Alternatively, the offset controlvalue 208 may be received from an external source. In one embodiment,the phase jumping mixer 221 is implemented in the same manner as thephase jumping mixer 217 so that, when the phase control word 232 matchesthe phase count value 230 (i.e., when the offset control value 208 iszero), the phase-jumping mixer 221 generates a mix clock signal 214having substantially the same phase as the mix clock signal 210generated within the tracking loop 203. The mix clock signal 214 isoutput to the clock tree circuit 223 which, in turn, generates multipleinstances of a device clock signal 216 (DCLK). In one embodiment, theclock tree circuit 219 within the tracking loop 203 is implemented inthe same manner as the clock tree circuit 223 within the offset clockgenerator 205 so that substantially equal delays are produced within theclock tree circuits 219 and 223. Accordingly, in the case of azero-valued offset control value 208, the feedback clock signal 212 anddevice clock signal 216 are substantially aligned in phase. In analternative embodiment in which multiple instances of the device clocksignal 216 are not required (e.g., the number of circuit elementsclocked by the device clock signal 216 is relatively small), the clocktree circuits 219 and 223 may be omitted.

Phase Jumping Mixer

FIG. 4 illustrates the phase jumping mixer 221 according to anembodiment of the invention (phase jumping mixer 217 may be implementedin the same manner as mixer 221). The mixer 221 includes a bank ofswitch-selectable differential amplifiers 249, mix logic 251, andamplifier biasing circuit 253. The amplifier bank 249 includes eightdifferential amplifiers, A, B, C, D, E and H, each formed by a pair oftransistors having drain terminals coupled respectively to a mix clockline 279 and complement mix clock line 281, and source terminals coupledin common to a corresponding one of eight switch elements 275. It shouldbe noted that the number of differential amplifiers (and switchelements) corresponds to the number of phase vectors generated by thereference loop (i.e., element 201 of FIG. 3) and therefore may be higheror lower in alternative embodiments. Also, in alternative embodimentsthe number of differential amplifiers may be different from the numberof switch elements (e.g., in an application in which the number of phasevectors is different from the number of differential amplifiers).

The mix clock line 279 and complement mix clock line 281 are pulled upto a predetermined reference voltage (supply voltage, VDD, in thisexample) by respective resistive elements, R. In the exemplaryembodiment of FIG. 4, the rightmost transistor within each differentialamplifier A-H is coupled to the mix clock line 279 and is referred toherein as the mix transistor, while the leftmost transistor is coupledto the complement mix clock line 281 and referred to as the complementmix transistor. Each of the mix transistors within the differentialamplifiers A-H is coupled to receive a respective one of eight phasevectors from a reference loop, thereby allowing each of the eight phasevectors to be selected to be mixed into the mix clock signal, MCLK andcomplement mix clock signal, /MCLK. Each of the complement mixtransistors is coupled to receive a phase vector that is the complementof the phase vector input to the corresponding mix transistor. By thisarrangement, whenever a given phase vector is selected to be mixed intothe mix clock signal, the complement phase vector is selected to bemixed into the complement mix clock signal.

In one embodiment, the switch elements 275 are transistor switches(e.g., MOS transistors) having control terminals coupled to receiverespective amplifier select signals, SA-SH, from the mix control logic251. When a given control signal, SA-SH, is asserted, the switchingtransistor coupled to receive the asserted signal is switched on,coupling the corresponding differential amplifier to the amplifierbiasing circuit 253. Switch elements 275 controlled by amplifier selectsignals SA-SD are coupled between differential amplifiers A-D,respectively, and an even-phase bias circuit 283 within the amplifierbiasing circuit 253, while switch elements 275 controlled by amplifierselect signals SE-SH are coupled between differential amplifiers E-H andan odd-phase bias circuit 285 within the amplifier biasing circuit 253.By this arrangement, a selected one of amplifier select signals SA-SDmay be asserted to enable a corresponding one of differential amplifiersA-D to contribute to generation of the mix clock signal and complementmix clock signal, while the others of the amplifier select signals aredeasserted to disable the corresponding differential amplifiers fromparticipating in the phase mixing operation. Similarly, a selected oneof amplifier select signals SE-SH may be switched on to enable acorresponding one of differential amplifiers E-H to contribute togeneration of the mix clock signal and complement mix clock signal,while the others of the amplifier select signals SE-SH are deasserted todisable the corresponding differential amplifiers.

The mix logic 251 receives the phase control word 232 (e.g., from adder235 of FIG. 3) and includes a range selector 261, bias word generator263, and bias word synchronizer 269. In one embodiment, the mostsignificant M bits of the phase control word 232 constitute a rangeselect value 258 (RSEL) and the remaining bits constitute aninterpolation weight 259 (IW). In the embodiment of FIG. 4, the rangeselect value 258 is a three-bit value in which each of the eightpossible bit patterns corresponds to one of eight phase ranges (i.e.,octants) bounded by a respective pair of phase-adjacent phase vectors.The range select value 258 is input to the range selector 261 whichdecodes the range select value 258 to generate the amplifier selectsignals, SA-SH. The following table illustrates the correspondencebetween the range select value 258 and amplifier select signals, SA-SH,in an exemplary embodiment of the range selector 261: TABLE 1 RSEL SA SBSC SD SE SF SG SH 000 1 0 0 0 1 0 0 0 001 0 0 1 0 1 0 0 0 010 0 0 1 0 00 1 0 011 0 1 0 0 0 0 1 0 100 0 1 0 0 0 1 0 0 101 0 0 0 1 0 1 0 0 110 00 0 1 0 0 0 1 111 1 0 0 0 0 0 0 1

FIG. 5 illustrates the correspondence between range select values, phasevectors and mix-enabled differential amplifiers (i.e., differentialamplifiers enabled to contribute to the mix clock signals, MCLK and/MCLK) in the mixer embodiment of FIG. 4. As an example, when the rangeselect value 258 is zero (000), amplifier select signals SA and SE areasserted (i.e., to a logic high state), thereby enabling amplifiers Aand E to contribute to generation of the mix clock signal and complementmix clock signal. In such a state, phase vectors t0 and t1 are mixedaccording to the bias currents drawn by biasing circuits 283 and 285 togenerate the mix clock signal, MCLK; and phase vectors /t0 and /t1 aremixed to generate the complement mix clock signal /MCLK. All otheramplifier select signals are driven low (as illustrated above in Table1), thereby disabling amplifiers B, C, D, F, G and H. Rotating throughthe selectable phase ranges, differential amplifiers E and C are enabled(and all others disabled) when the range select value 258 is 1 (001),selecting phase vectors t1 and t2 to be mixed to generate the mix clocksignal, and vectors /t1 and /t2 to be mixed to generate the complementmix clock signal; differential amplifiers C and G are enabled when therange select value 258 is 2 (010), selecting phase vectors t2 and t3 tobe mixed to generate the mix clock signal and phase vectors /t2 and /t3to be mixed to generate the complement mix clock signal; differentialamplifiers G and B are enabled when the range select value 258 is 3(011), selecting phase vectors t3 and /t0 to be mixed to generate themix clock signal, and phase vectors /t3 and t0 to be mixed to generatethe complement mix clock signal; differential amplifiers B and F areenabled when the range select value 258 is 4 (100), selecting phasevectors /t0 and /t1 to be mixed to generate the mix clock signal, andphase vectors t0 and t1 to be mixed to generate the complement mix clocksignal; differential amplifiers F and D are enabled when the rangeselect value 258 is 5 (101), selecting phase vectors /t1 and /t2 to bemixed to generate the mix clock signal, and phase vectors t1 and t2 tobe mixed to generate the complement mix clock signal; differentialamplifiers D and H are enabled when the range select value 258 is 6(110), selecting phase vectors /t2 and /t3 to be mixed to generate themix clock signal, and phase vectors t2 and t3 to be mixed to generatethe complement mix clock signal; and differential amplifiers H and A areenabled when the range select value 258 is 7 (111), selecting phasevectors /t3 and t0 to be mixed to generate the mix clock signal, andphase vectors t3 and /0 to be mixed to generate the complement mix clocksignal.

Reflecting on the phase diagram of FIG. 5, it can be recognized that foreach selectable phase range, an even numbered phase vector (i.e., t0,t2, /t0, or /t2) is mixed with an odd numbered phase vector (i.e., t1,t3, /t1, or /t3). Thus, referring again to FIG. 4, because all but aselected one of the even-phase differential amplifiers (A, B, C, D) isdecoupled from the amplifier biasing circuit 253 at a given time (i.e.,by opening selected switches 275) and all but one of the odd-phasedifferential amplifiers (E, F, G, H) is decoupled from the amplifierbiasing circuit 253 at a given time, the total number of component biascircuits required within the amplifier biasing circuit 253 is reduced bya factor of four. That is, because only one of the even-phasedifferential amplifiers is selected at a time, a single even-phase biascircuit may be shared by the four even-phase differential amplifiers.Similarly, a single odd-phase bias circuit may be shared by the fourodd-phase differential amplifiers. As discussed below, each of thecomponent bias circuits 283, 285 within the amplifier biasing circuit253 is formed by multiple, digitally controlled biasing transistors andtherefore is substantially larger than the single-transistor biasingcircuits used in the prior-art arrangement of FIG. 2 (i.e., transistors153). Thus, the sharing of component bias circuits 283, 285 amongmultiple differential amplifiers A-D and E-H significantly reduces thetotal area consumed by the amplifier biasing circuit 253, making thedigitally controlled, multi-transistor biasing circuits more feasiblethan if a dedicated component biasing circuit was required for eachdifferential amplifier within amplifier bank 249.

Referring again to the mix logic 251, the interpolation weight 259 isinput to an inverter 262 to generate a complement interpolation weight,with both the interpolation weight and complement interpolation weightbeing supplied to respective inputs of multiplexers 265 and 267. In oneembodiment, the complement interpolation weight is used to bias adifferential amplifier coupled to receive the leading phase vector for aselected phase range, and therefore constitutes a leading-vectorinterpolation weight, LVI. Conversely, the uncomplemented interpolationweight is used to bias a differential amplifier coupled to receive thetrailing phase vector for the selected phase range, and thereforeconstitutes a trailing-vector interpolation weight, TVI. The selectionof trailing- and leading-vector interpolation weights is made by themultiplexers 265, 267 in response to the least significant bit (LSB) ofthe range select signal, which indicates whether the leading phasevector for a selected phase range is an even or odd phase vector. Thus,when the trailing phase vector for a selected phase range is an oddphase vector (i.e., the range select value 258 is 0, 2, 4 or 6), TVI isoutput by multiplexer 265 as an odd-phase bias control word, OBC, andLVI is output by multiplexer 267 as an even-phase bias control word,EBC. Conversely, when the trailing phase vector for a selected phaserange is an even phase vector (i.e., the range select value 258 is 1, 3,5 or 7), TVI is output by multiplexer 267 as the even-phase bias controlword, EBC, and LVI is output by multiplexer 265 as the odd-phase biascontrol word, OBC. By this arrangement, as the interpolation weight 259is incremented from zero to a maximum value, interpolation is shiftedfrom the leading phase vector to the trailing phase vector,interpolation being shifted entirely to the trailing phase vector whenthe interpolation weight 259 reaches a maximum value (i.e., LVI=0 sothat the leading phase vector does not contribute to the pahse of themix clock signal). When the interpolation weight rolls over from amaximum value to zero (i.e., in response an increment indication insignal 206, or an increment in the offset control value 208), a newphase range is selected, with the trailing phase vector for thepreceding phase range becoming the leading phase vector for the newphase range. By the operation of the multiplexers 265 and 267, theinterpolation weight applied to the new leading phase vector is themaximum-valued LVI; the same weight as previously applied as a trailingphase vector weight (maximum-valued TVI). Accordingly, mixing progressessmoothly through the transition between adjacent phase ranges.

The selection of trailing and leading-vector interpolation weights asthe even- and odd-phase bias control words for the embodiment of FIG. 4is illustrated by the following table: TABLE 2 RSEL OBC EBC 000 TVI LVI001 LVI TVI 010 TVI LVI 011 LVI TVI 100 TVI LVI 101 LVI TVI 110 TVI LVI111 LVI TVI

The even- and odd-phase bias control words, EBC and OBC, are strobedinto respective storage registers 271 and 273 within the bias wordsynchronizer 269 in response to the device clock signal 216 (other clockor strobe signals may be used to strobe the bias control words into thebias word synchronizer 269 in alternative embodiments). The even-phasebias control word is output from storage register 271 to the even-phasebias circuit 283 within the amplifier biasing circuit 253, and theodd-phase bias control word is output from storage register 273 to theodd-phase bias circuit 285 within the amplifier biasing circuit 253. Inthe embodiment of FIG. 4, the even- and odd-phase bias circuits 283 and285 are implemented by digitally controlled current sources(current-sinking digital-to-analog converters referred to herein ascurrent DACs) in which multiple digitally controlled, current sinkingtransistors are coupled in parallel with one another and in seriesbetween a switch-selected differential amplifier and a reference voltage(ground in this example). By this arrangement, each current sinkingtransistor within the even- and odd-phase current DACs 283 and 285 maybe quickly switched on or off according to the constituent bits of theeven- and odd-phase bias control words, thereby rapidly increasing ordecreasing the bias current applied to a selected differential amplifierand producing a corresponding rapid phase change in the mix clock signaland complement mix clock signal. Thus, in contrast to the prior-artarrangement of FIG. 2 in which changes in DAC-generated bias voltagesrequire significant time (e.g., multiple device clock cycles) to settleat the gates of biasing transistors 153, the current sinking transistorswithin the even- and odd-phase current DACs 283, 285 are rapidlyswitched between on and off states, without need to wait for gatevoltages to settle at precise levels.

The bias word synchronizer 269 ensures that transitions of constituentbits within the even- and odd-phase bias control words, EBC and OBC, areapplied to the constituent current sinking transistors within the even-and odd-phase current DACs 283 and 285 at substantially the same time,significantly reducing the phase jitter that otherwise may result fromtiming differences in the generation of the bias control words (e.g.,due to propagation of the trailing-vector interpolation word throughinverter 262). Also, because each current sinking transistor within theeven- and odd-phase current DACs 283 and 285 is switched fully on or off(i.e., in contrast to the analog bias voltages used to set a precisetransconductance value for transistors 153), the current DACs aresubstantially less susceptible to control line noise (i.e., noise on thebias control word paths 284 and 286) than in the prior-art arrangementof FIG. 2. Accordingly, the capacitive elements used to reducenoise-induced jitter in the prior-art arrangement of FIG. 2 may beomitted in the embodiment of FIG. 4, enabling more rapid transitionbetween successive bias control words and therefore further speedingphase transitions in the mix clock signal and complement mix clocksignal. The electrical lengths of the bias control word paths 284 and286 may also be equalized (e.g., through layout symmetry, or capacitiveor inductive compensation on one or both of the paths 284 and 286) tofurther reduce difference between arrival times of the even- andodd-phase bias control words within the even- and odd-phase current DACs283 and 285.

Slew Control

FIG. 6 illustrates the even- and odd-phase current DACs 283 and 285 ofFIG. 4 according to an embodiment of the invention. The even- andodd-phase current DACs have identical structures and each include Nbiasing transistors (i.e., current sinking transistors) coupled toreceive a respective, constituent bit of a bias control word (i.e.,EBC[i] or OBC[i], i being an integer between 0 and N−1). Each of thebiasing transistors 303 ₀-303 _(N−1) within current DAC 283 is coupledin series between a selected even-phase differential amplifier (i.e., A,B, C or D) and a reference voltage (ground in this example). Similarly,each of the biasing transistors 305 ₀-305 _(N−1) within current DAC 285is coupled in series between a selected odd-phase differential amplifier(i.e., E, F, G or H) and the reference voltage. Thus, when switched on,the digitally-controlled biasing transistors 303 and 305 draw currentdirectly from a selected differential amplifier and thereby enable thephase vectors applied at the control terminals of the differentialamplifier to affect the level of the mix clock lines 279 and 281.Accordingly, the current DACs 283 and 285 are referred to herein asin-line current DACs to emphasize the digitally controlled switching ofbiasing transistors coupled in series with the differential amplifierswithin the amplifier bank 249 of FIG. 4.

In one embodiment, each of the N biasing transistors 303 within currentDAC 283 (and transistors 305 within current DAC 285) is binary weighted(e.g., by transistor sizing) such that, when switched on, biasingtransistor 303 ₀ draws a reference current, I_(REF), (indicated by thedesignation “x1” in FIG. 6), biasing transistor 303 ₁ draws I_(REF)x2(i.e., x2 weighting), biasing transistor 303 ₂ draws I_(REF)x4, and soforth to biasing transistor 303 _(N−1) which, when switched on, drawsI_(REF)x2 ^(N−1) (i.e., x2 ^(N−1) weighting). Bits 0 to N−1 of theeven-phase bias control word (i.e., EBC[0] to EBC[N−1]) are supplied tothe gates of biasing transistors 303 ₀-303 _(N−1),respectively. By thisarrangement, the total current drawn by the biasing transistors 303₀-303 _(N−1) ranges from zero to I_(REF)x(2 ^(N)−1) according to thecorresponding value of the even-phase bias control word, EBC[N−1:0].Similarly, the total current drawn by the biasing transistors 305 ₀-305_(N−1) ranges from zero to I_(REF)x(2 ^(N)−1) according to thecorresponding value of the odd-phase bias control word, OBC[N−1:0].

Still referring to FIG. 6, each of binary weighted transistors 307 ₀-307_(N−1) is coupled in series with a respective one of theparallel-coupled biasing transistors 303 ₀-303 _(N−1) and is biased by asteady state bias voltage, nBIAS, to establish the reference currentI_(REF). Each of binary weighted transistors 309 ₀-309 _(N−1) issimilarly coupled in series with a respective one of theparallel-coupled biasing transistors 305 ₀-305 _(N−1) and biased by thenBIAS voltage. In one embodiment, each of transistors 307 ₀-307 _(N−1)and 309 ₀-307 _(N−1) is biased to draw a current substantially equal toits weight (i.e., x1, x2, x4, etc . . . ) multiplied by I_(REF) suchthat the total current drawn by the current DACs 283 and 285 may belinearly stepped in I_(REF) increments in response to correspondingincrements in the even- and odd-phase bias control words.

The nBIAS voltage level is set such that, when all the biasingtransistors within a given one of current DACs 283 and 285 are switchedon and all the biasing transistors within the other of the current DACsare switched off, the resulting mix clock signal slews at a desiredrate. In one embodiment, the nBIAS voltage is set to an empiricallydetermined level by a current mirroring circuit. In an alternativeembodiment, described in greater detail below, the nBIAS voltage isgenerated within a reference loop (e.g., element 201 of FIG. 3) andreflects the voltage level required to achieve a desired slew ratewithin a delay element of the reference loop for a given voltage,temperature and process.

Because the even- and odd-phase bias control words are complements ofone another (i.e., due to the operation of inverter 262 of FIG. 4), whena biasing transistor 303 of a given weight is turned on within theeven-phase current DAC 283, a corresponding transistor 305 (i.e.,transistor having the same weight) within the odd-phase current DAC 285is turned off, and vice-versa. This circumstance is used to advantage inan alternative amplifier biasing circuit embodiment depicted in FIG. 7.As shown, each of the even-phase and odd-phase biasing transistors 303and 305 of a given weight are coupled differentially to one another andin series with a reference-current-setting transistor 311 of similarweight. That is, EBC[i] and OBC[i] (i representing an integer between 0and N−1) are supplied to respective gate terminals of transistors 303_(i) and 305 _(i), each of which is coupled in series between areference-current-setting transistor 311 _(i) and a respective set ofdifferential amplifiers (i.e., even phase differential amplifiers A-D orodd-phase differential amplifiers E-H). Thus, a single set oftransistors 311 is used in place of the two sets of transistors 307 and309 of FIG. 6 to establish a reference current for both even-phase andodd-phase biasing transistors 303 and 305. By this arrangement, space issaved and any distortion due to operational differences betweentransistors 307 and 309 (e.g., due to temperature gradient) is avoided.Steady-state bias voltage, nBIAS is supplied via line 312 to the gateterminals of the reference-current-setting transistors 311 to establishthe desired slew rate in the mix clock signal. Capacitive element 313(or a distribution of capacitive elements) may be coupled to line 312 toreduce noise.

FIG. 8 illustrates the phase steps produced in a mix clock signal as theinterpolation weight generated by the mix logic 251 of FIG. 4 isincremented from zero to a maximum value. Initially, when theinterpolation weight is zero, the trailing-vector interpolation weight,TVI, is at a minimum value (e.g., zero), and the leading-vectorinterpolation weight, LVI, is at a maximum value. Thus, the differentialamplifier coupled to receive the leading phase vector is biased at fullscale and the differential amplifier coupled to receive the trailingphase vector is biased at zero. Consequently, the resultant mix clocksignal has a phase according to the leading phase vector and is slewedat a maximum rate determined by the current flowing through theleading-vector differential amplifier. Referring briefly to FIG. 4, itcan be seen that the slew rate of the mix clock signal, dv/dt, is afunction of the capacitance value of the capacitive elements (C) coupledto the mix clock lines, and the full-scale current drawn by theamplifier biasing circuit. That is, dv/dt=I_(DAC-FULLSCALE)/C. Thus, thefull-scale current drawn by the amplifier biasing circuit 253, a valuecontrolled by nBIAS, may be increased or decreased to produce acorresponding increase or decrease in the maximum slew rate of the mixclock signal.

When the interpolation weight is incremented, TVI is incremented and LVIdecremented so that, before the trailing vector begins to transition (atime 334 indicated by the 45° line), the slew rate of the mix clocksignal is incrementally lower than the maximum slew rate. That is, thecurrent flowing through the differential amplifier is incrementallylower than the maximum current, resulting in a proportional reduction indv/dt. At time 334, when the trailing vector begins to transition, themix clock slew rate increases to the maximum slew rate (i.e., the sum ofcurrents drawn by the even-phase current DAC and odd-phase current DACequal the full-scale biasing current). Consequently, the mix clocksignal, having slewed at a slightly reduced rate, crosses the midpointvoltage 332 slightly later than under the previous LVI/TVI condition,thereby achieving a stepped phase delay relative toleading-vector-driven phase 331. Thus, by incrementally reducing LVI andincreasing TVI, the midpoint crossing of the mix clock signal isincrementally stepped from a leading-vector-aligned phase to atrailing-vector-aligned phase.

As discussed in reference to FIG. 3, an offset control value 208 isadded to the phase count value 230 in adder 235 to establish a desiredphase offset between the device clock signal 216 and the feedback clocksignal 212. Because the device clock signal 216 is generated by an openloop circuit (i.e., the offset clock generator), the accuracy of thephase offset is dependent on the linearity of the mixing operationperformed within the phase-jumping mixers 217 and 221. That is, anynonlinearity in the mixing operation is manifested as unequal phasesteps within a given phase range, thereby producing potential phaseerror in the mix clock signal.

Referring to FIG. 8, it can be seen that a general requirement formixing linearity is that the mix clock signal, when slewing at a maximumrate, not cross the mid point voltage 332 before the trailing vectorbegins to transition. Otherwise, as illustrated by phase step diagram360, the contribution of the trailing vector to the mix clock slew rate(illustrated by 363) will be inconsequential for all LVI-TVI weightingsin which the leading vector 361 alone produces a midpoint crossing(i.e., crossing the voltage indicated by 332) prior to the 45° time.Thus, as shown by the unequal phase steps in phase step diagram 360 andby the dashed line 353 in phase angle plot 350, distorted, non-linearmixing of the leading and trailing phase vectors is produced when themaximum mix clock slew rate is too high; a distortion referred to hereinas S-curve distortion. By contrast, so long as the mix clock signal,when slewing at a maximum rate, crosses the midpoint voltage 332 when orafter the trailing vector begins to transition, the phase steps will besubstantially equal as shown in phase step diagram 358, producing thelinear mix curve 351 shown in phase angle plot 350. If the mix clocksignal slews too slowly, the overall signal swing may not reach thedesired maximum and minimum voltage levels. That is, the peak-to-peakvoltage of the signal swing, V_(SWING), is reduced. Accordingly, bysetting the full-scale DAC current (i.e., within the amplifier biasingcircuit 253 of FIG. 4) such that the mix clock signal slews to themidpoint voltage 332 in a time substantially equal to the mix clockperiod divided by the number of selectable phase ranges, S-curvedistortion may be avoided without undue reduction of V_(SWING).Expressed analytically, the mix clock signal slew rate is set to(V_(SWING)/2)/(T_(MCLK)/#Vectors), where V_(SWING) is the desiredpeak-to-peak amplitude of the mix clock signal, T_(MCLK) is the periodof the mix clock signal, and #Vectors is the number of phase vectorsused to subdivide the mix clock period.

Clock Signal Swing

Referring again to FIG. 4, it should be noted that any adjustment in thebias currents drawn by the amplifier biasing circuit 253 will produce acorresponding change in the minimum voltage level of the mix clocksignal. That is, an increase in the biasing current will produce anincreased voltage drop across the resistive elements, R, and a decreasein the biasing current will produce a corresponding decrease in thevoltage drop. This is generally undesirable as changes in the mix clocksignal swing may have disruptive consequences in downstream circuits,for example, causing distortion in conversion from small swing tocomplementary-MOS signaling levels.

FIG. 9 illustrates a mixer embodiment 375 in which the resistance valuesof mix clock line pull-up elements 381 and 383 are dynamically adjustedby a bias voltage, pBLAS, to maintain a relatively constant mix clocksignal swing over changes in bias current drawn by the amplifier biasingcircuit 253. Thus, the bias current drawn by the amplifier biasingcircuit 253 may be increased or decreased as necessary to maintain adesired mix clock slew rate, and the resistance of the resistiveelements 381 and 383 correspondingly decreased or increased to maintaina relatively constant mix clock voltage swing. That is,dv/dt=(M₁xI_(DAC-FULLSCALE))/C, andV_(MIN SWING)=(M₁X_(DAC-FULLSCALE))×(R/M2), where M₁ is adjusted bynBIAS andM_(2 is adjusted by pBLAS. By maintaining a substantially constant proportionality between M1 and M2 (i.e., M1/M2=K), the mix clock voltage swing is maintained at a substantially constant value over changes in the mix clock slew rate.)

Reflecting on the effect of the nBIAS and pBLAS voltages within thephase jumping mixer 375, it can be seen that the nBIAS voltageconstitutes a clock slew control signal and the pBIAS voltageconstitutes a clock amplitude control signal. Referring briefly to FIG.3, in one embodiment of the invention, the nBIAS and pBLAS voltages aregenerated within the reference loop 201 and output to the phase jumpingmixers 217 and 221 as the slew rate control signal 226 and the amplitudecontrol signal 228, respectively.

FIG. 10 illustrates a reference loop for generating phase vectors t0-t3,/t0-/t3, and the nBIAS and pBIAS voltages that are output to the phasejumping mixers as control signals 226 and 228. The reference loopincludes a series of delay elements4150-^(4154, a phase detector 403, control word generator 405, and bias voltage generator 407. Each of the delay elements 415 receives a complementary pair of input clock signals and generates a corresponding pair of complementary phase vectors. The nBIAS voltage is supplied to each delay element 415 via bias line 410 to control the slew rate of the phase vectors generated by the delay element, and therefore the overall phase delay achieved by the delay element. The pBIAS voltage is supplied to each delay element 415 via bias line 412 to control the amplitude of the output clock for the delay element.)

In the reference loop 401 of FIG. 10, five delay elements 415 ₀-415 ₄are provided, each being biased to generate phase vectors thattransition from a peak voltage level (maximum or minimum) to the swingmidpoint voltage (i.e., (V_(PEAK-MAX)+V_(PEAK-MIN))/2) over a timeinterval that corresponds to 45 degrees of the cycle time of referenceclock signal 202. By this arrangement, each delay element 415 generatesa pair of phase vectors that are delayed by 45 degrees (i.e., of thephase vector cycle time) relative to the input phase vectors (or, in thecase of delay element 415 ₀, relative to the input reference clocksignals 202, 204). Thus, delay element 415 ₀ receives the complementaryreference clock signals 202 and 204 and outputs phase vectors t0 and /t0in response. Delay element 415 ₁ receives phase vectors t0 and /t0 andoutputs phase vectors t1 and /t1 to delay element 415 ₂, which outputsphase vectors t2 and /t2 to delay element 415 ₃, which outputs phasevectors t3 and /t3 to delay element 415 ₄ which outputs phase vectors t4and /t4. Phase vectors t4 and t0 are input to the phase detector 403which outputs a signal 404 having a high or low state according to whichphase vector leads the other. If phase vector t4 leads phase vector t0,then the total delay through delay elements 415 ₁-415 ₄ is less than afull cycle of the reference clock signal 202 and therefore is too short.Conversely, if phase vector t0 leads phase vector t4, then the delaythough delay elements 415 ₁-415 ₄ is more than a full cycle of thereference cock signal 202 and therefore is too long. If t4 leads t0(delay too short), the phase detector 403 outputs a decrement signal(e.g., a low-state signal 404) to the control word generator 405 whichresponds by decrementing a slew control word 406. If t4 lags to (delaytoo long), the phase detector outputs an increment signal (e.g., ahigh-state signal 404) to the control word generator 405 whichincrements the slew control word 406 in response. The slew control word406 is output to the bias voltage generator 407. The bias voltagegenerator 407 outputs the nBIAS and pBLAS voltages on lines 410 and 412,respectively, according to the slew control word 406 from the controlword generator 405. In one embodiment, the bias voltage generator 407includes a frequency-dependent bias control circuit to adjust the nBIASand pBLAS voltages according to the frequency of the reference clocksignal 202, thereby allowing the reference loop 401 to be operated overa relatively broad range of reference clock frequencies.

FIG. 11 illustrates an exemplary embodiment of a delay element 415 thatmay be used within the reference loop of FIG. 10. The delay element 415includes a differential amplifier formed by differentially coupledtransistors 443, resistive pull up elements 447 and 449, capacitiveelements 455 and 457, and current source 445. Differential clock signals(CLK_(IN) and /CLK_(IN)) are applied to the gate terminals oftransistors 443, such that, when CLK_(IN) is high and /CLK_(IN) is low,all or substantially all of the current drawn by current source 445flows through resistive element 447, thereby pulling down complementoutput clock line 450 and enabling output clock line 452 to charge.Conversely, when /CLK_(IN) is high and CLK_(IN) is low, substantiallyall of the current drawn by current source 445 flows through resistiveelement 449, thereby pulling down output clock line 452 and enablingcomplement output clock line 450 to charge. The rate at which the clocklines 450 and 452 are pulled down is proportional to the current, I,drawn by current source 445 (i.e., dv/dt=I/C, C being one of capacitiveelements 455 and 457) which, in turn, is controlled by the nBIASvoltage. Thus, the nBIAS voltage controls the slew rate of thecomplementary output clock signals (CLK_(OUT) and /CLK_(OUT)) generatedon clock lines 450 an 452, and therefore the clock phase delay achievedby the delay element 415. In the exemplary embodiment of FIG. 10, thenBIAS voltage is adjusted through the closed loop operation of thereference loop 401 until the slew rate of the output clock signal,CLK_(OUT), produces a midpoint crossing after a time periodsubstantially equal to T_(REFCLK)/8; a 45 degree phase delay. If thephase delay is greater than 45 degrees, phase vector t4 will lag phasevector to, causing nBIAS to be increased, thereby increasing the outputclock slew rate and decreasing the phase delay. Conversely, if the phasedelay is less than 45 degrees, phase vector t4 will lead phase vectort0, causing nBIAS to be decreased, thereby decreasing the output clockslew rate and increasing the phase delay. Because the nBIAS voltage isgenerated through the closed loop operation of the reference loop, nBIASis adjusted as necessary to maintain the desired output clock slew rateover gradual changes in voltage and temperature (i.e., environmentalchanges), and over process variations from device to device.

Still referring to FIG. 11, the pBIAS voltage is used to adjust theresistance of resistive elements 447 and 449 as necessary to maintain asubstantially constant output clock amplitude (i.e., V_(SWING)) over agiven range of nBIAS voltages. In one embodiment, the pBIAS and nBIASvoltages are generated in a manner that maintains a substantiallyconstant proportionality between the voltages so that, like nBIAS, pBIASis adjusted in response to changes in process, voltage and temperatureto maintain a desired output clock amplitude. Thus, the nBIAS and pBIASvoltages constitute slew rate and amplitude control signals,respectively, that are adjusted by a closed loop circuit to compensatefor changes in process voltage and temperature (PVT).

Comparing the delay element of FIG. 11 to a given one of thedifferential amplifiers A-H of FIG. 4, it can be seen that thedifferential amplifier and delay element have essentially the samestructure. Accordingly, by using the PVT-compensated nBIAS voltage toestablish the mix bias current within the phase-jumping mixers, adesired mix clock slew rate is maintained through changes in process,voltage and temperature. Similarly, by using the PVT-compensated pBLASvoltage to establish the resistance of the mix clock pull-up elements,R, within the phase-jumping mixer 221, a desired mix clock amplitude ismaintained through changes in process, voltage and temperature.

FIG. 12 illustrates an exemplary embodiment of the voltage generationcircuit 489 of FIG. 10 and its interconnection to an exemplary delayelement 485. The voltage generation circuit 487 includes a dual-controlcurrent source 491 that includes component current sources 493 and 495that draw bias currents, K₂I_(SCB) and K₁I_(DAC), respectively. Asdiscussed below, current source 493 is controlled by aswitched-capacitor biasing circuit that generates a bias control valueaccording to the frequency of the reference clock signal. Current source495 is controlled by the slew control word 406 from the control wordgenerator 405 of FIG. 10 and therefore enables digital adjustment of thedelay through the delay element 487 to achieve a desired slew rate inoutput clock signals, CLK_(OUT) and /CLK_(OUT).

Still referring to FIG. 12, the current drawn by dual-control currentsource 491 (i.e., I_(X)=K₁I_(DAC)+K₂I_(SCB)) is used to establish thepBIAS voltage at the gate of diode-configured transistor 497. Transistor497 is coupled in a current-mirroring configuration with transistor 499and, via line 412, with transistors 503 and 505 of the delay element485. In one embodiment, transistor 499 is substantially identical (i.e.,same length-width ratio) to transistor 497 so that current I_(X) flowsthrough transistor 499 thereby establishing the nBIAS voltage at thegate of diode-configured transistor 501. As shown, the nBIAS voltage isapplied via line 410 to the gate of biasing transistor 511 within thedelay element 485 to achieve a bias current, I_(nBIAS), equal to (orsubstantially equal to) K₃I_(X)=K₃(K₁I_(DAC)+K₂I_(SCB)), K3 being aconstant established, for example, by the width ratios betweentransistors 511 and 501 (and/or by the width ratios between transistors497 and 499).

The pBIAS voltage applied to transistors 503 and 505 increases inproportion to the nBIAS voltage applied to the gate of biasingtransistor 511, and is used to control the resistive load presented bytransistors 503 and 505. Because the current flow through transistors503 and 505 is substantially proportional to the gate-to-source voltage,V_(GS), the resistive load presented by transistors 503 and 505 issubstantially inversely proportional to the pBIAS voltage and thereforeis inversely proportional to the nBIAS voltage. That is, the resistiveload presented by transistors 503 and 505 is inversely proportional tothe current drawn by biasing transistor 511 and therefore is decreasedas the current drawn by biasing transistor 511 is increased. Because theoutput clock signals developed on lines 450 and 452 swings approximatelybetween V_(MAX)=V_(DD) and V_(MIN)=V_(DD)−(R₅₀₃×I_(nBIAS)), the inverseproportionality between R₅₀₃ (i.e., the resistance presented bytransistor 503) and I_(nBIAS) serves to maintain V_(MIN) (and thereforeV_(SWING)) at a relatively constant level as I_(nBIAS) is adjusted toachieve a desired slew rate. Additional resistive elements (e.g., diodeconfigured transistors, transistors biased at predetermined operatingpoints, etc.) may additionally be coupled to the output clock lines 450and 452 to provide a baseline resistance which is adjusted by changes inthe resistive values of transistors 503 and 505.

FIG. 13 illustrates an embodiment of the dual-control current source 491of FIG. 12. The dual-control source 491 includes the frequency-trackingcurrent source 493 and the digitally controlled current source 495described in reference to FIG. 12. The frequency-tracking current source493 includes a pair of transistors 539 and 541 coupled in series betweena supply voltage and the output 542 of a follower-configured amplifier545. A capacitive element 543 is coupled between ground and the junctionof transistors 539 and 541 (i.e., to the drain terminal of transistor539 and the source terminal of transistor 541). The complementaryreference clock signals 202 and 204 are input to gate terminals of thetransistors 539 and 541, respectively, such that, when clock signal 202is high (and clock signal 204 is low), transistor 539 is switched on tocharge the capacitive element 543, and transistor 541 is switched off.The voltage output by the follower-configured amplifier 545 is set to avalue (determined by reference voltage, V_(REF)) lower than V_(DD) sothat, when the clock signal 204 is high (and clock signal 202 is low),transistor 541 is switched on to discharge the capacitor throughdiode-configured transistor 547, thereby generating a bias voltage atthe gate of transistor 547 according to the discharge current. Thecurrent flowing through the series coupled transistors 539 and 541 (andtherefore through diode-configured transistor 547), I_(SCB), is(V_(DD)−V_(REF))/Z, where Z is 1/(CxF_(CLK)). Note that the transistors539 and 541 contribute a resistive component to the impedance, but aredominated by the CxF_(CLK) term. Thus, I_(SCB) is substantially equal to[(V_(DD)−V_(REF))×C]×F_(CLK), and, because V_(DD), V_(REF) and C arerelatively constant, is therefore proportional to the frequency of thereference clock signal. This is a desirable result as the requiredoutput clock slew rate in the delay elements and the phase-jumpingmixers increases linearly with the frequency of the reference clocksignal. Diode-configured transistor 547 is coupled in a current mirrorconfiguration with transistor 551, such that a current K2xI_(SCB) flowsthrough transistor 551, the K₂ term being a constant established by therelative length-width ratios of transistors 551 and 547.

The digitally controlled current source 495 includes a current DAC 521coupled to receive the slew control word (SCW) from the control wordgenerator (i.e., element 405 of FIG. 10). In one embodiment, the currentDAC 521 includes N binary weighted transistors (designated x1, x2, x4, .. . , x2 ^(N−1) in FIG. 13) coupled in parallel with one another, eachhaving a gate terminal coupled to receive a respective bit of the slewcontrol word. By this arrangement, the current drawn by DAC 521,I_(DAC), is adjustable between zero to 2^(N−1)xI U in steps of I_(UNIT)(I_(UNIT) being the current drawn by the x1 transistor, when switchedon) according to the value of the slew control word. I_(DAC) flowsthrough diode-configured transistor 531 which is coupled in a currentmirror configuration with transistor 533. In one embodiment, transistors531 and 533 are substantially the same size so that, by virtue of thecurrent mirror, I_(DAC) flows through transistor 533 and thereforethrough diode configured transistor 535. Transistor 535 is coupled in acurrent mirror configuration with transistor 553 so that currentK₁xI_(DAC) flows through transistor 553. The multiplier K₁ is a constantdetermined by the relative length-width ratios of transistors 553 and535. In an alternative embodiment, transistors 533 and 531 may be usedto establish K₁ instead of (or in addition to) transistors 535 and 553.

Reflecting on the operation of the dual-control current source 491, itshould be noted that, for a reference clock signal having a givenfrequency, the current drawn by the frequency-tracking current source493, K₂xI_(SCB), is substantially constant. Consequently, the adjustablerange of the current source 491 extends from a mini-mum value,K₂xI_(SCB), when I_(DAC) is zero; to a maximum value,K₂xI_(SCB)+K₁I_(DAC), when I_(DAC) is at full-scale. In one embodiment,illustrated in FIG. 14, the multiplier K₂ is selected such that anominally mid-point I_(nBIAS) value (576) may be achieved in a fastprocess comer (i.e., fabrication process that yields the highestacceptable transconductance value for a transistor of a given size) whenI_(DAC) is set to zero, and that may also be achieved in a slow processcomer (i.e., fabrication process that yields the lowest acceptabletransconductance value for a transistor of a given size) when I_(DAC) isset to full-scale. That is, as shown in FIG. 14, both the slow processline (SLOW) and the fast process line (FAST) cross the mid-pointI_(nBIAS) line 576 at opposite extremes of the slew control word, with anominal process having an adjustable I_(nBIAS) range centered around themid-point I_(nBIAS) 576. I_(nBIAS) for the fast process corner rangesfrom the mid-point I_(nBIAS) 576, to the mid-point I_(nBIAS) plusK₁I_(DAC) (designated “Imax(F)” in FIG. 14 ). I_(nBIAS) for the slowprocess comer ranges from the mid-point I_(nBIAS) 576 to the mid-pointI_(nBIAS) minus K₁xI_(DAC) (designated “Imin(S)” in FIG. 14).

Although the phase-jumping locked loop architecture of FIG. 3 andcomponent circuits thereof described in reference to FIGS. 4-14 havebeen described as generating a device clock signal having the samefrequency as a reference clock signal (i.e., a delay-locked loop), thephase-jumping locked loop architecture and component circuits mayreadily be adapted to form a phase-locked loop (PLL) circuit in whichthe output clock signal is a frequency multiple of the reference clocksignal. Referring to FIG. 3, for example, clock divider circuitry may beused within the reference loop 201 to generate frequency-multipliedphase vectors, and the frequency-multiplied phase vectors mixed withinthe phase-jumping mixers 217 and 221 as described above. In either typeof phase-jumping locked loop circuit, DLL or PLL, the process-,temperature- and voltage-tracking bias voltages used to control the slewrate and amplitude of clock signals within the reference loop may beoutput to the phase-jumping mixers 217 and 221 to maintain substantiallylinear mixing and substantially constant output clock signal swing overchanges in process, temperature and voltage.

System Application of Phase Jumping Locked Loop Circuit

FIG. 15 illustrates a system 700 in which a phase jumping locked loopcircuit 709 (i.e., a DLL or PLL circuit) according to embodimentsdescribed above in reference to FIGS. 3-14 may be used. The system 700may be used, for example, within a computing device (e.g., mobile,desktop or larger computer), networking equipment (e.g., switch, router,etc.), consumer electronics device (e.g., telephone, camera, personaldigital assistant (PDA), etc.), or any other type of device in which aPLL or DLL circuit may be used. More specifically, the system 700 may bea memory subsystem or any other subsystem within such computing device,networking equipment, consumer electronics device, etc.

The system 700 includes a pair of integrated circuits (ICs) 701 and 703coupled to one another via a transmit signal path 702 and a receivesignal path 704. In the embodiment, shown, the signal paths 702 and 704are unidirectional high-speed serial links for conducting serializedtransmissions from one IC to the other. In alternative embodiments,either or both of the links may be bi-directional (i.e., withappropriate circuitry provided to select which of the ICs is enabled totransmit on the link at a given time), and multiples of such signalpaths may be provided to enable transmission of parallel groups of bits(e.g., each group of bits forming a data or control word (e.g., command,address, etc.) or portion of a data or control packet). Also, thetransmit signal path 702, receive signal path 704, and/or sharedtransmit-receive signal path may be a multi-drop bus that is coupled toadditional ICs. The ICs 701 and 703 may be peers (e.g., each IC iscapable of independently initiating a signal transmission to the other),or master and slave. Also, the relative status of the ICs 701 and 703may change from time-to-time such that one IC is a master at a firsttime, then a slave at another time, and/or a peer at another time.

IC 701 is shown in simplified block diagram form and includes a transmitcircuit 711, receive circuit 713, locked loop circuit 709, andapplication logic 715. As shown, the locked loop circuit 709 is coupledto receive complementary reference clock signals, CLK and /CLK, from anoff-chip reference clock generator 705, and outputs a phase-locked clocksignal 706 to the transmit circuit 711 and the receive circuit 713. Inan alternative embodiment, the reference clock signals, CLK and /CLK,may be generated within IC 701 or IC 703. A configuration circuit 717(e.g., register, one-time programmable circuit, non-volatile memory,etc.) may be included within the application logic 715 to store one ormore offset control values (OFFSET) that are used to establish a phaseoffset between clock signal 706 and reference clock signal, CLK. Notethat clock signal 706 may include a complementary pair of clock signalsas described above. Also, while the locked loop 709 is depicted asproviding a clock signal to both the transmit circuit 711 (i.e., atransmit clock signal) and to the receive circuit 713 (i.e., a samplingclock signal), separate locked loop circuits may be provided to generateseparate transmit and sampling clock signals. Alternatively, multipleclock generation circuits may be provided within the locked loop circuit709 to generate separate transmit and sampling clock signals. Forexample, in an embodiment in which locked loop 709 is a DLL circuitimplemented as shown in FIG. 3, an additional phase jumping mixer andclock tree circuit may be provided to generate a transmit clock inresponse to a separate offset control value (OFFSET). Also, although twoICs are depicted in FIG. 15 (i.e., ICs 701 and 703), the circuits withineach of the ICs may alternatively be implemented in a single IC (e.g.,in a system-on-chip or similar application), with signal paths 702 and704 being routed via metal layers or other signal conducting structuresfabricated within the IC. Also, if distinct ICs are used as shown inFIG. 15, the ICs may be packaged in separate IC packages (e.g., plasticor ceramic encapsulation, bare die package, etc.) or in a single ICpackage (e.g., multi-chip module, paper thin package (PTP), etc.).

Phase Jumping

Because the phase-jumping mixer of the present invention exhibitsrelatively fast settling time between phase steps (i.e., as compared tothe prior-art phase mixer described in reference to FIG. 2), a number ofapplications which require rapid, relatively large, phase changes becomepossible. For example, in an application in which the phase jumpingmixer is used to generate a transmit clock (i.e., to time transmissionof signals), it may be desirable to select a different phase offsetbetween the transmit clock and a reference clock signal according to adata (and clock) propagation distance. More specifically, a respectiveoffset control value may be established for each recipient device in asignaling system and selected (e.g., from a lookup table or othermemory) by a transmitting device according to which recipient device isthe intended recipient of an outgoing transmission. Because thetransmitting device may need to transmit to one or more differentrecipient devices in rapid succession, delay in generating transmitclock signals having the desired phase offsets would present asubstantial bottleneck in such a system. Using the phase jumping mixerof FIG. 4 (or FIG. 9), a sequence of transmit clocks having differentphase offsets may be rapidly generated by changing the offset controlvalue 208. This type of operation is referred to herein asdestination-based phase jumping. While some settling time in theresultant mix clock signal is still necessary, the settling time is, ingeneral, substantially shorter than in the prior art mixer describedabove.

Embodiments of the present invention may also be used to achieve a rapidsuccession of different phase alignments of a sampling clock signal,with each different phase alignment corresponding to a respectivetransmission source within a signaling system. For example, in amaster/slave system in which slave transmissions to a master deviceoccur deterministically in response to master device commands orrequests (i.e., when the master device issues a command or request, themaster device may anticipate a responsive transmission from the slave apredetermined time later), the master device may select a previouslydetermined sampling clock offset according to which slave device isscheduled to transmit at a given time. Such operation is referred toherein as source-based phase jumping and may be implemented within amaster device, for example, by storing a respective offset control valuefor each slave device in a signaling system and selecting (e.g., from alookup table or other memory) different ones of the stored offsetcontrolled values according to the identities of the slave devicesscheduled to transmit. More generally, source-based phase jumping may beused in any device that has or receives forehand information oftransmission sources. Such forehand information may result from systemdeterminism (i.e., predetermined response times to system events such ascommands, requests, interrupts, timing signals, etc.) or from othercommunications including, without limitation, communications viaout-of-band signaling channels (e.g., handshaking signals).

Both destination and source-based phase jumping may be implementedwithin the same integrated circuit device (e.g., one or more masterdevices within a master/slave system) and a shared memory structure usedto store offset control values for the various transmission destinationsand sources. Offset control values may be determined, for example, bytransmission of predetermined test patterns between system devices tolearn the leading and lagging phase boundaries beyond which transmissionerrors are detected. Methods and apparatuses for performing such timingcalibration operations are disclosed, along with other locked-loopapplications in which embodiments of the present invention may be used,in U.S. patent application Ser. No. 09/421,073, filed Oct. 19, 1999(entitled “Bus System Optimization”), and U.S. Pat. No. 6,321,282, eachof which is hereby incorporated by reference in its entirety.

Embodiments of the present invention may also be applied to achievedestination and/or source-based phase jumping in signaling systems inwhich integrated circuit devices (or components within an integratedcircuit device) are clocked by different clock signals having nominallythe same frequency, but arbitrary phase relationships. In such a system,the phase offsets between various transmit and sampling clock signalsused to time signal transmission and reception may be systematicallydetermined and used to generate offset control values. The offsetcontrol values may then be dynamically retrieved (e.g., from a lookuptable or other memory) to time data reception and/or transmissionoperations in one or more of the integrated circuits according to thesource or destination of the data transmission. In one such system, forexample, a memory controller mounted to a motherboard (or othersubstrate) is coupled to multiple memory modules (i.e., daughterboardsor other substrates having one or more integrated circuit memory devicesmounted thereon) via a common signaling path, the memory modules beingclocked by respective clock signals having nominally the same frequency,but arbitrary phase relationships. The memory controller may receive anyor all of the clock signals provided to the memory modules, and/or aseparate clock signal. In one embodiment, the memory controller includesa DLL circuit or PLL circuit according to the present invention toachieve rapid, source and/or destination-based phase jumping.

The phase jumping capability of the phase jumping mixer of FIGS. 4 (orthe phase jumping mixer of FIG. 9) may also be applied in phase lockingoperations performed upon device power up or exit from a reduced powerstate. In phase locking operations, the phase of a sampling clock signaland/or transmit clock signal generated by a locked loop circuit (i.e.,DLL or PLL) is incrementally rotated through a cycle (or part of acycle) of a reference clock signal until a desired phase offset isreached. Referring to FIG. 3, the phase jumping mixers within trackingloop and offset clock generator the may be used to jump betweendifferent phases in search of a desired phase offset. For example, thephase jumping mixer 217 of FIG. 3 may be used to rapidly jump betweentarget phases of the feedback clock signal 212 in a binary search inwhich the search range is repeatedly halved to converge on a phase countvalue that establishes phase alignment between the reference clocksignal 202 and the feedback clock signal. Similarly, the phase jumpingmixer 221 of FIG. 3 may be used to rapidly jump between a number ofphase offsets in an effort to locate leading and trailing edges of adata valid window within an incoming data signal (i.e., leading andtrailing edges of a data eye). In both cases, a combination of binaryand linear searching techniques may be used, for example, by using abinary search to reduce the search range to a predetermined phase range,then stepping through the predetermined phase range in linear incrementsuntil a desired phase offset is reached. In general, any searchingtechnique in which it is desirable to rapidly switch between differentphase offsets may potentially benefit from the short settling time ofthe phase jumping mixers of the invention.

Time-Multiplexed TX/RX Clock Line

FIG. 16 illustrates a signaling device 745 in which transmit and receivephase control values are alternatively supplied to a phase jumping mixer751 such that a corresponding transmit clock signal and receive clocksignal are alternatively output onto a shared clock line 754. That is,the clock signal line 754 is effectively time-multiplexed so that,during a data reception interval, an incoming data waveform on signalpath 778 is sampled by a receive circuit 775 in response to transitionsof a receive clock signal generated on the clock line 754 and, during adata transmission interval, data is transmitted onto the signal path 778by a transmit circuit 773 in response to transitions of a transmit clocksignal generated on the clock line 754. Note that clock line 754 may begated within the clock tree circuit 753 or elsewhere such that thesignal transitions used to time the reception and transmission of dataconstitute strobe signals rather than clock signals. Also, signal path778 may be a single-ended or differential signal path.

The signaling device 745 includes a tracking loop 747, offset clockgenerator 749, transceiver 781 and application logic 771. The signalingdevice additionally includes a reference loop supply a set of phasevectors 758 (and optionally, the slew control signal 226 and amplitudecontrol signal 228 described in reference to FIG. 3) to the trackingloop 747 and the offset clock generator 749. The tracking loop 747includes a phase counter 757, phase jumping mixer 751, clock tree 753and phase detector 759 that operate generally in the same manner as thecorresponding components described in reference to FIG. 3 to generate aphase count value 756. The phase count value 756 represents a phaseoffset between the reference clock signal 760 and a reference phasevector of phase vectors 758 (i.e., one of phase vectors 758 arbitrarilyselected to represent a 0 degree phase angle). Within the tracking loop747, the phase count value 756 constitutes a phase control word that issupplied to the phase jumping mixer 751 to produce a feedback clocksignal 752 that is phase aligned with the reference clock signal 760.

The offset clock generator 749 includes a pair of storage circuits 761and 765, summing circuit 750, select circuit 769, phase jumping mixer751 and clock tree 753. The storage circuits 761 and 765 are used tostore a transmit phase offset value and a receive phase offset value,respectively, which are output to the summing circuit 750. In theembodiment of FIG. 16, the summing circuit 750 includes a pair of addercircuits 763 and 767 each of which is coupled to receive the phase countvalue 756 from the tracking loop. Adder circuit 763 sums the phase countvalue 756 with the transmit phase offset value from register 761 togenerate a transmit phase control word 764, and adder circuit 767 sumsthe phase count value 756 with the receive phase offset value fromregister 765 to generate a receive phase control word 766. The phasecontrol words 764 and 766 are input to the select circuit 769 whichoutputs a selected one of the phase control words (i.e., as selectedphase control word 770) to the phase jumping mixer 751 according to thestate of a direction signal 768 received from the application logic 771.The phase jumping mixer 751 mixes a selected pair of phase vectors 758according to the selected phase control word 770 to produce a mix clocksignal 772. The mix clock signal propagates through the clock treecircuit 753 to generate an output clock signal on clock line 754, theoutput clock signal having a phase offset relative to the referenceclock signal 760 according to the selected phase control word 770. Thus,when the direction signal 768 from the application logic 771 indicates areceive operation, the receive phase control word 766 is selected togenerate a receive clock signal on clock line 754, the receive clocksignal being used to time the sampling instant within a receive circuit775 (i.e., within transceiver 781), and the data samples captured inresponse to the receive clock signal being supplied to the applicationlogic 771 via receive data line 776. Conversely, when the directionsignal 768 from the application logic 771 indicates a transmitoperation, the transmit phase control word 764 is selected to generate atransmit clock signal on clock signal line 754, the transmit clocksignal being used to time the transmission of data on signal path 778 bythe transmit circuit 773, the data being supplied to the transmitcircuit 773 by application logic 771 via transmit data line 774.

The application logic 771 includes circuitry specific to the generalfunction of the integrated circuit device (e.g., memory controller,memory device, processor, application-specific integrated circuit(ASIC), programmable logic device (PLD), or any other type of integratedcircuit device. In one embodiment, the transmit circuit 773 is apull-down type transmit circuit that pulls signal path 778 down from aprecharged level (e.g., pulled up to a supply voltage) to transmit asymbol other than a zero-valued symbol, and that transmits a zero-valuedsymbol by allowing the signal path 778 to remain at the prechargedlevel. Thus, the application logic 771 may effectively disabletransmitter 773 from affecting the state of the signal path 778 during adata reception interval by outputting a value onto the transmit dataline 774 that corresponds to a zero-valued symbol. In an alternativeembodiment in which the transmitter 773 is a push-pull output driver (orany other type of output driver that affects the state of the signalpath 778 regardless of the value of the symbol being transmitted), theapplication logic may output a disable signal to the transmitter 773 todisable the transmitter from affecting the state of the signal path 778during a data reception interval. Also, while the transmit phase offsetvalue and the receive phase offset value are depicted as being stored indedicated storage circuits 761 and 765, a shared storage circuit (e.g.,a multi-entry memory array) may alternatively be used.

FIG. 17 illustrates the relationship between an exemplary data waveformon signal path 778, the direction signal 768 and correspondingtransitions of the transmit and receive clock signals generated on clockline 754. When the direction signal 768 is in a first state (low in thisexample), a transmit clock signal is generated on clock signal line 754and used to time the transmission of successive data values on signalpath 778. In the embodiment shown, a data value (e.g., a symbolrepresented by one of at least two discrete signal levels, or more thantwo discrete signal levels in a multi-level signaling embodiment) istransmitted on signal path 778 in response to each transition of thetransmit clock signal (i.e., a double data-rate signal in which eachsymbol is transmitted during a successive half-cycle of the transmitclock signal). In alternative embodiments, more or fewer data values maybe transmitted during each cycle of the transmit clock signal. As shownat 784, the application logic 771 transitions the direction signal fromlow to high in anticipation of receiving data via the signal path 778.During a turnaround interval shown at 785, no data is transmitted orreceived on the signal path 778, thereby allowing the signal path 778 tosettle. The duration of the turnaround interval 785, T_(TURNAROUND), maybe longer or shorter than the time between successive data transmissionsand/or data receptions. The offset clock generator 749 of FIG. 16responds to the transition of the direction signal 768 by selecting thereceive phase control value 766 to be supplied to the phase jumpingmixer 751 and therefore transitioning the phase of the clock signalgenerated on clock line 754 from the transmit clock signal phase to thereceive clock signal phase. Thus, after the turnaround interval 785, theclock signal on clock line 754 has the desired phase offset for samplingdata within receive circuit 775 and is used to sample data values fortransfer to the application logic 771.

FIG. 18 illustrates an alternative circuit arrangement for generatingthe phase control value 770 within the offset clock generator 749 ofFIG. 16. As shown, a select circuit 801 is coupled to receive thetransmit and receive offset values from the storage circuits 761 and765, respectively. The select circuit outputs a selected one of thephase offset values (i.e., selected according to the state of thedirection signal 768) to a summing circuit implemented by an adder 803.The adder 803 sums the selected phase offset value with the phase countvalue 756 received from the tracking loop 747 to generate the phasecontrol value 770. The phase control value 770 is supplied to the phasejumping mixer 751 and used to generate an output clock signal asdescribed in reference to FIG. 16. Note that the registers 761 and 765depicted in FIGS. 16 and 18 may alternatively be implemented by a memoryarray having at least two storage entries (i.e., to store the transmitand receive phase offset values), an address decoder to select betweenthe storage entries in response to an address signal (e.g., thedirection signal 768) and an output port (e.g., bit lines coupled tocolumns of storage elements within the memory array) to supply thecontent of the selected storage entry to the adder 803.

Reflecting on the operation of the device of FIG. 16, it should be notedthat, absent the fast phase jumping ability of the mixer 751 within theoffset clock generator 749, a transmit or receive clock signal wouldlikely require a time significantly longer than the turnaround intervalto stabilize on the clock signal line 754. Thus, the fast phase jumpingability of the mixer 751 enables generation of both transmit and receiveclock signals on the same clock signal line, avoiding the need for anadditional phase mixer and clock tree. More generally, the architectureof device 745 may be used in any application in which it is desirable toquickly transition an output clock signal between two or more phaseoffsets. Also, while the phase jumping mixer 751 may be implemented bythe phase jumping mixers described above in reference to FIGS. 4-14, anycircuit capable of rapidly transitioning the phase of an output clocksignal according to the selection between two or more phase controlvalues may alternatively be used within the clock generating circuit inplace of the phase jumping mixer 751.

Phase Searching

To save power during periods of non-communication in a high-speedsignaling system, delay locked loop and phase locked loop circuits areoften disabled from tracking a reference clock signal (the referenceclock signal itself being shut off in some systems). Beforecommunication is restored in such systems, the locked loop circuits arere-enabled in a wake-up operation. In many systems, the time required tocomplete the wake-up operation is the dominant factor in how quicklycommunication may be restored, and is directly related to the timerequired for the locked loop circuit to regain phase lock with thereference clock signal.

FIG. 19 illustrates a binary phase searching operation in which phasejumping within the tracking loop 203 of the locked loop circuit of FIG.3 (or tracking loop 747 of FIG. 16) is used to reduce the time requiredto regain phase lock within the locked loop circuit 200. At the start ofa wake-up operation, when the locked loop circuit is enabled (e.g., byenabling the reference clock signal 202 to transition), the trackingloop generates an initial feedback clock signal, F0, having an arbitraryphase with respect to the reference clock signal (REF CLK). The initialfeedback clock signal may be generated based on a previously generated(and now stale) phase count value or, in the case of initial wake-up(i.e., at device power-up), a random phase count value or a phase countvalue that has been reset to a predetermined value (e.g., zero).

FIG. 20 illustrates possible phase relationships between the referenceclock signal (REF CLK) and the feedback clock signal (FCLK). If a risingedge transition 818 of the feedback clock signal falls within a highinterval 819 of the reference clock signal, the feedback clock signalwill be determined by a phase detector (i.e., element 247 of FIG. 3 or757 of FIG. 16) to lag the reference clock signal. Conversely, if arising edge transition 820 of the feedback clock signal falls within alow interval 821 of the reference clock signal, the feedback clocksignal will be determined by the phase detector to lead the referenceclock signal. Thus, shortly after a wake-up operation is begun, thephase detector 247 of FIG. 3 (or phase detector 757 of FIG. 16) willoutput a phase adjust signal that indicates whether the feedback clocksignal leads or lags the reference clock signal.

Reflecting on FIG. 20, it can be seen that if the feedback clock signalis indicated to lag the reference clock signal, the feedback clocksignal lags the reference clock signal by at most 180°. Conversely, afeedback clock signal indicated to lead the reference clock signal doesso by at most 180 degrees. Thus, as shown in FIG. 19, the initiallead/lag indication by the phase detector may be used to halve aninitial 360° search range, SR₀ (i.e., range of possible phase offsetsbetween the initial feedback clock signal and reference clock signal),thereby producing 180° search range, SR₁. Accordingly, by transitioningthe phase of the feedback clock signal (i.e., in a phase-jumpingoperation) to a phase angle in the center of search range SR₁, andrepeating the lead/lag determination for the new feedback clock signal(F1), search range SR₁ may be halved to produce search range, SR₂.Search range SR₃ may similarly be determined by jumping to feedbackclock signal F2 (i.e., in the center of search range SR₂) and halvingsearch range SR₂ based on the subsequent lead/lag determination. Searchrange SR₃ is similarly halved to produce search range SR₄ based on thelead/lag determination for feedback clock phase F3. This operation iscontinued with the size of the phase jump being halved for eachsuccessively determined search range, until the desired phase offset isdetermined or until the size of the phase jump reaches a minimum value.

FIG. 21 illustrates a tracking loop 823 for performing the phasesearching operation illustrated in FIG. 19. The tracking loop 823includes a phase detector 759, phase counter 827, phase jumping mixer751 and clock tree 753, all of which operate generally as described inreference to FIG. 16 and FIG. 3 to generate a feedback clock signal 752.The tracking loop 823 additionally includes search control logic 825 andadder circuit 829 which are used in the phase search operation to loadthe phase counter 827 with a sequence of conditionally-determined phasecount values.

When a locked loop circuit which includes the tracking loop 823 isenabled (e.g., in a wake-up operation), the value in the phase counter827 may be stale, random or otherwise may not reflect the phasedifference between the reference clock signal 760 and the feedback clocksignal 752. Accordingly, the phase of the feedback clock signal 752 mayhave any phase offset relative to the reference clock signal 760 and thephase search operation of FIG. 19 is undertaken to achieve a phase countvalue 756 within the phase counter 827 that produces phase alignmentbetween the reference and feedback clock signals 760 and 752.

Referring to FIGS. 21 and 22, initially, at block 851, the searchcontrol logic 825 deasserts enable signal 826 to disable the phasecounter 827 from incrementing and decrementing the phase count 756 inresponse to the phase adjust signal 824 (IJ/D) from the phase detector759. The search control logic 825 also outputs a digital value thatrepresents a phase jump angle; the jump angle initially being set to avalue that corresponds to one-fourth of a full cycle of the referenceclock signal 760 (i.e., 360°/4=(maximum phase count+1)/4). Note thatdifferent initial jump angles may be used, for example, in systems orapplications in which the overall search range is less than a full cycleof the reference clock signal 760. The search control logic 825 receivesone or more lead/lag indications 824 from the phase detector (e.g.,having vote logic to determine a lead/lag result according to whethermore lead indications than lag indications, or vice versa, are receivedwithin a given time interval) and thereby determines, at decision block855 whether the feedback clock signal 752 leads or lags the referenceclock signal 760. If the feedback clock signal 752 leads the referenceclock signal 760, the search control logic 825 outputs a positive jumpangle to adder 829 (i.e., via path 830), which responds by adding jumpangle to the present phase count value 756 to produce an updated phasecount value on path 832. The updated phase count value is loaded intothe phase counter 827 in response to assertion of a load signal 828 bythe search control logic 825. Thus, as illustrated at block 859 of FIG.22, the search control logic 825 responds to the lead determination at855 by loading the phase counter with a sum of the current phase countvalue and the jump angle, thereby retarding the phase of the feedbackclock signal 752 by a phase angle that corresponds to the jump angle.If, at decision block 855, the feedback clock signal 752 is determinedto lag the reference clock signal 760, then the search control logic 825outputs a negative jump angle to the adder 829 (e.g., by operation of acircuit within the search control logic 825 that changes the sign of thejump angle in response to a lag indication), thereby effecting asubtraction of the jump angle from the current phase count value andadvancing the phase of the feedback clock signal 752 by a phase anglethat corresponds to the jump angle. At block 861 of FIG. 22, the searchcontrol logic 825 compares the jump angle to a minimum value. If thejump angle is less than the minimum value, then the search operation iscompleted and the phase counter 827 is re-enabled at 865 (i.e., searchcontrol logic 825 asserts the enable signal 826), thereby enablinglinear, incremental phase tracking within the phase counter 827 inresponse to the phase adjust signal 824 from the phase detector 759. Ifthe jump angle is not less than the minimum value, then the jump angleis halved at block 863 and a new iteration of the binary searchoperation is begun at 855. In one embodiment, the search control logic825 includes a shift register to halve the jump angle by right-shiftinga binary representation of the jump angle by one bit.

Searching for Leading and Trailing Edges of a Data Eye

After phase lock is achieved within the tracking loop of a phase jumpinglocked loop circuit, another phase search may be performed in the offsetclock generator (i.e., element 749 of FIG. 16, or 205 of FIG. 3) todetermine the phase offset between a desired sampling instant and thereference clock signal. Because the tracking loop generates a phasecount value that represents an offset between a reference phase vectorand the reference clock signal, determining the phase offset for thedesired sampling instant may be achieved by determining an offset valueto be added to the phase count value to produce a receive clock signal(i.e., sampling clock signal) having the desired phase offset from thefeedback clock signal. In one embodiment, this operation involvesinitiating a data transmission in a remote device to produce an incomingtest data stream, then adjusting the phase of the receive clock signalto determine pass-fail phase boundaries that correspond to leading andtrailing edges within data eyes of the incoming data stream. The desiredsampling instant may then be selected at the midpoint between thepass-fail phase boundaries.

While the task of determining pass-fail phase boundaries may be achievedby incrementing a phase offset value (e.g., the value 208 of FIG. 3, orthe value stored in register 765 of FIG. 16) in unitary steps, andtesting for correct reception of the test data at each step, thisoperation can take considerable time, extending the overall systeminitialization and/or wake-up time. In one embodiment of the invention,the fast-phase jumping ability of the phase jumping locked loop of FIG.3 (or FIG. 16) is employed to perform a coarse linear search for leadingand trailing edges within data eyes of the test data sequence, forexample, by phase jumping through a sequence of clock signals, referredto herein as search vectors, that are offset from one another by a phaseangle smaller than an expected minimum eye width. By this operation, atleast one of the search vectors, referred to herein as a pass-vector,should fall within the incoming data eye and therefore yield properreception of the test data sequence. Accordingly, a leading edge of thedata eye is known to have a phase offset between a fail-vector (i.e.,search vector which fails to yield proper reception of the test datasequence), and an immediately succeeding pass-vector. Similarly, atrailing edge of the data eye is known to have a phase offset between apass-vector and an immediately succeeding fail vector. The fail-vectorsand pass-vectors which bound the leading and trailing edges of theincoming data eye may then be used as bounds in a binary search torapidly locate the edges of the data eye. A linear search (or other typeof search) may be used to locate the edges of the data eye instead of orin addition to the bounded binary search (e.g., bounded binary search toreduce the search range, followed by linear search to determine aprecise phase offset). The overall effect of the coarse linear searchfollowed by fine search (binary, linear and/or other), is tosignificantly reduce the number of phase offsets that are evaluated tolocate the edges (and therefore the midpoint) of the data eye,potentially producing a corresponding reduction in the amount of timerequired to determine the desired receive clock phase offset.

As shown in FIG. 23, the phase offset of the incoming data eye 876 mayhave any phase offset within a cycle time of the feedback clock signal,but should have at least some minimum eye width 875. In one embodiment,the minimum eye width is a specified value that is used to determine anumber of coarse search ranges by dividing an offset that corresponds toa full cycle angle of the feedback clock signal (i.e., 360°=max offsetvalue+1) by the angle that corresponds to the minimum eye width 875. Forexample, if the duration of the minimum eye width 875 corresponds to 75°of the feedback clock cycle time, the number of search ranges would be360°/75°=4 (plus a remainder). In one embodiment, the integer number ofsearch ranges is increased by one to ensure a coarse search range thatis smaller than the phase angle of the minimum eye width 875. That is,the number of coarse search ranges=[360°/(phase angle of minimum eyewidth)]+1. Other formulations for determining the number of coarsesearch ranges may be used in alternative embodiments. =p FIG. 24illustrates the division of a cycle of the feedback clock signal (andtherefore the reference clock signal) into five search ranges, SR0-SR4,in response to a minimum eye width having an exemplary phase angle of75°. The minimum eye width may correspond to a substantially smaller orlarger phase angle in alternative embodiments. As an example, an actualeye 877 is depicted in FIG. 24 as extending through most of search rangeSR3 and into part of search range SR4. A search vector is generated foreach of the search ranges, SR0-SR4, in sequence by phase jumping from aninitial phase offset of zero (search vector, SV₀) through a sequence ofphase offsets that correspond to the phase angle of the search ranges.That is, a digital phase jump value that corresponds to the size of eachsearch range (i.e., (max phase count +1)/# search ranges) iscumulatively added to the offset control value 208 of FIG. 3 (or thereceive clock phase offset value stored in register 765 of FIG. 16) toproduce the sequence of search vectors, SV₀-SV₄, that correspond to thecenter points of search ranges SR0-SR4, respectively. Thus, in theexample of FIG. 24, search vectors SV₀, SV₁, SV₂ and SV₄ fall outsidethe data eye 877 and therefore constitute fail-vectors, while searchvector SV₃ falls within the data eye and therefore constitutes apass-vector (i.e., search vector SV₃ will yield correct data reception;search vectors SV₀-SV₂ and SV₄ will not). Accordingly, a leading edge ofthe data eye is bounded by search vectors SV₂ and SV₃, while a trailingedge of the data eye 877 is bounded by search vectors SV₃ and SV₄. Thesebounding vectors may now be used as outer limits in subsequent,finer-granularity searches for the leading and trailing edges of thedata eye 877.

FIG. 25 is a flow diagram of a coarse linear search for leading andtrailing edges of a data eye according to an embodiment of theinvention. At 901, the number of search ranges is determined asdescribed above according to the minimum eye size; a search vectoroffset (SVO), which represents a phase offset value summed with thephase count value from a tracking loop to generate a given searchvector, is initialized to zero; a jump angle value (JMP ANGLE) isinitialized as described above according to the number of search ranges;Boolean variables, LE_(FOUND), LE_(SEARCH), TE_(FOUND) and TE_(SEARCH),used to indicate the status of the leading and trailing edge searchesare initialized to indicate a false state (FALSE); and phase offsetvariables, EYE LE_(FAIL), EYE LE_(PASS), EYE TE_(PASS) and EYETE_(FAIL), used to store the phase offsets of search vectors determinedto bound the leading and trailing edges of the data eye are initializedto the value of the search vector offset (zero in this example). Notethat in alternative embodiments, the number of search ranges may be apredetermined value (i.e., an initial value) or may be generated usingother formulations. The jump angle may also be a predetermined value inalternative embodiments.

At 903, the search vector offset is loaded into the offset register toproduce an initial search vector (i.e., clock signal generated by theoffset clock generator 205 of FIG. 3 or 749 of FIG. 16) that issubstantially phase aligned with the feedback clock signal. A patterntransfer test is executed at 905 (i.e., receiving a predeterminedsequence of test data values transmitted by a remote device). If thetest data sequence was not properly received (i.e., pass/faildetermination at 907), then the current search vector is a fail-vectorand the search operation branches to 909. If the test data sequence wasproperly received, the current search vector is a pass-vector, andsearch operation branches to 917. In the case of a fail-vector, if aleading edge of the data eye has not been found (i.e., any precedingexecutions of the pattern transfer test have not yielded a fail-vectordetermination followed by a pass-vector determination), then thefail-vector represents a possible bounding vector for a leading edge ofthe data eye. Accordingly, at 911, Boolean value LE_(SEARCH) is set totrue to indicate detection of a fail-vector, and the search vectoroffset is recorded in EYE LE_(FAIL), a value that represents the phaseangle of the bounding fail-vector for a leading edge of the data eye. At913, Boolean value TE_(SEARCH) is inspected to determine whether apass-vector was located in a prior execution of the pattern transfertest at 905. If so, the fail-vector detected in the present iterationconstitutes an outer bound of a trailing edge of the data eye.Accordingly, at 915, Boolean value TE_(FOUND) is set to TRUE to indicatethat pass- and fail-vectors that bound the trailing edge of the data eyehave been found, and the search vector offset that yielded the presentfail-vector is recorded in EYE TE_(FAIL). Also, Boolean valueTE_(SEARCH) is set to FALSE to prevent further update to the EYETE_(FAIL) value.

Returning to 907, in the case of a pass-vector, if a trailing edge ofthe data eye has not been found (i.e., any preceding executions of thepattern transfer test have not yielded a pass-vector determinationfollowed by a trail-vector determination), then the pass-vectorrepresents a possible bounding vector for a trailing edge of the dataeye. Accordingly, at 919, Boolean value TE_(SEARCH) is set to true toindicate detection of a pass-vector, and the search vector offset isrecorded in EYE TE_(PASS), a value that represents the phase angle ofthe bounding pass-vector for a trailing edge of the data eye. At 921,Boolean value LE_(SEARCH) is inspected to determine whether afail-vector was located in a prior execution of the pattern transfertest 905. If so, then the pass-vector detected in the present iterationconstitutes an outer bound of a leading edge of the data eye.Accordingly, at 923, Boolean value LE_(FOUND) is set to TRUE to indicatethat fail- and pass-vectors that bound the leading edge of the data eyehave been found, and the search vector offset that yielded the presentpass-vector is recorded in EYE LE_(PASS). Also, Boolean valueLE_(SEARCH) is set to FALSE to prevent further update to the EYELE_(PASS) value.

After fail-vector processing in blocks 909-915 or pass-vector processingin blocks 917-923, the jump angle is summed with the search vectoroffset at 925 to produce a search vector offset that corresponds to thenext search vector. At 927, the search vector offset is compared with amaximum value to determine whether all the search vectors have beenevaluated. If so, the coarse linear search is completed and a binaryedge search is executed at 931. The binary edge search is described infurther detail below in reference to FIG. 26. In one embodiment, even ifall the search vectors have not been evaluated, the coarse linear searchmay still be concluded if the leading and trailing edges of the data eyehave been found (i.e., LE_(FOUND) and TE_(FOUND) are determined to betrue in 929). Otherwise, the coarse linear search is repeated, startingat 903, for the updated search vector offset.

The following table illustrates the result of a coarse linear searchaccording to FIG. 25 assuming the data eye location depicted in FIG. 24:Test SVO Result LE_(SRCH) LE_(FOUND) LE_(FAIL) LE_(PASS) TE_(SRCH)TE_(FOUND) TE_(PASS) TE_(FAIL) 0° Fail True False 0° 0° False False 0°0° 72° Fail True False 72° 0° False False 0° 0° 144° Fail True False144° 0° False False 0° 0° 216° Pass True True 144° 216° True False 216°0° 288° Fail True True 144° 216° True True 216° 288°Thus, at the conclusion of the coarse linear search, a leading edge ofthe data eye has been determined to be bounded by phase offsets of 144°and 216°, and a trailing edge of the data eye has been determined to bebounded by phase offsets of 216° and 288°. Note that the search vectoroffset is a digital value, but is listed in degrees in the table abovefor purposes of illustration.

Still referring to FIG. 25, in one embodiment, if none of the searchvector offsets yields a pass-vector determination, the size of the jumpangle is decreased (e.g., by a predetermined factor or by a fixedamount), and the coarse linear search repeated. This shrinking of thejump angle produces a corresponding reduction in the size of the searchranges, and may be repeated until at least one pass-vector is found.Similarly, if no fail-vector is identified, the size of the jump anglemay be decreased and the coarse search repeated until at least onefail-vector is found. In such embodiments, the initial size of thesearch range (or minimum eye width) need not be specified, as the systemwill iteratively shrink or expand the search ranges (i.e., by jump angledecrease or increase) until pass-fail boundaries are located.

FIG. 26 illustrates a bounded binary search that may be executed tolocate the phase offset of a leading edge of the data eye afterexecution of the coarse linear search of FIG. 25. Initially, at 935,bounding variables B1 and B2 are loaded with the phase offset values(EYE LE_(FAIL) and EYE LE_(PASS), respectively) determined in the coarselinear search to bound the phase offset of the leading edge of the dataeye. At 937, the jump angle (i.e., digital value used to establish aphase jump size) is assigned a value equal to half the size of the phaserange defined by bounding phase offsets (i.e., (B2−B1)/2). At 939, a sumof the leading bounding variable, B1, and the jump angle (i.e., B1+JMPANGLE) is loaded into the offset register (i.e., to control the phaseoffset of the clock signal generated by the offset clock generator 205of FIG. 3 or 749 of FIG. 16) to produce an initial binary search vectorthat falls substantially midway between the phase vectors represented bybounding variables B1 and B2. A pattern transfer test is executed at 941by receiving a predetermined sequence of test data values transmitted bya remote device and comparing the sequence test data values with anexpected sequence. If the test data sequence was not properly received(i.e., test determined not to have passed at 943), then the searchvector falls outside the data eye and the bounding variable B1 is loadedwith the offset of the current search vector (i.e., B1+JMP ANGLE) at947, thereby moving the fail-vector offset (represented by B1) closer tothe leading edge of the data eye and halving the search range. If thepattern transfer test is determined to have passed at 943, then thebounding variable B2 is loaded with the offset of the current searchvalue at 945, thereby moving the pass-vector offset (represented by B2)closer to the leading edge of the data eye and halving the search range.At 949 the jump angle is halved in preparation for the next iteration ofthe bounded binary search. At 951 the jump angle is compared with aminimum jump angle (which may be, for example, a programmable value). Ifthe jump angle is less than the minimum jump angle, then the boundedbinary search is concluded and a stepwise linear search is optionallyperformed at 953 to find the precise phase offset of the leading edge ofthe data eye (i.e., the precise pass-fail boundary). If the jump angleis not less than the minimum jump angle, then the bounded binary searchis iterated with the smaller jump angle, starting at 939.

Upon conclusion of the bounded binary search at 951 (and optionally thelinear search at 953), the bounded binary search may be repeated todetermine the phase offset of the trailing edge of the data eye; thebounding variables B1 and B2 being assigned trailing edge boundingoffsets (EYE TE_(PASS) and EYE TE_(FAIL) values, respectively), insteadof the leading edge bounding offsets shown in 935; and the operations in947 and 945 being swapped to account for the opposite direction of thetransition between pass- and fail-vectors.

As discussed above, fine linear searches may be used to determine theprecise phase offsets of leading and trailing edges of a data eyeinstead of bounded binary searches. In one embodiment, a coarse linearsearch is performed as described in reference to FIG. 25 to locate apair of phase offsets that bound a leading edge of the data eye and apair of phase offsets that bound a trailing edge of the data eye, then afine linear search is performed within the phase range bounded by eachpair of phase offsets to determine the precise phase offsets of theleading and trailing edges of the data eye. In such an embodiment, thetotal number of phase search operations (i.e., phase change plus phasecomparison) performed may be expressed as follows:

(1) N=C+F₁+F₂, where C is the number of phase search operationsperformed in the coarse linear search; F₁ is the number of phase searchoperations performed in a fine linear search (i.e., stepwise incrementof phase control value, rather than a discontinuous jump) for the phaseoffset of the leading edge of the data eye, and F₂ is the number ofphase search operations performed in a fine linear search for thetrailing edge of the data eye. Letting M represent the total number ofselectable phase offsets within the complete searchable range, then F₁and F₂ may be expressed as follows:F ₁ =F ₂=(M−C)/C.   (2)

For example, if an 9-bit phase control value is used to control thephase offset of the mix clock signal generated by a phase jumping mixer,and twenty coarse phase search operations are performed to locate thebounding phases of the leading and trailing edges of the data eye, thenM=2⁹=512, and F₁=F₂=(512−20)/20=25 (i.e., after rounding up from 24.6 toan integer value). Thus, 25 stepwise phase search operations areperformed between bounding coarse phase offsets to locate the leadingedge of the data eye and another 25 stepwise phase comparison operationsare performed between bounding coarse phase offsets to locate thetrailing edge of the data eye, yielding a total of N=20+25+25=70 phasesearch operations to precisely locate the phase offsets of the leadingand trailing edges of the data eye.

Substituting the right-hand side of expression (2) for the F₁ and F₂terms in expression (1), the following expression for N is obtained:N=C+2(M−C)/C.   (3)In locked loop circuits for which M is a predetermined-value andtherefore fixed (M may alternatively be a programmable or adjustablevalue), it can be seen that N is a nonlinear function of C. Based onthis insight, expression (3) may be rewritten as a differentialexpression and solved for a relative minima (the second derivative ofexpression (3) is positive for C>0, so that the zero-valued firstderivative is a relative minima) as follows:N=C+2MC ⁻¹−2 {rewriting expression (3) to simplify thedifferential}  (4)dN/dC=1−2MC ⁻²   (5)0=1−2MC ⁻² {setting the differential to zero to solve for the relativeminima}  (6)C=(2M)^(1/2)   (7)Thus, for a locked loop circuit having M selectable phase offsets withina searchable range, the number of coarse linear search operations, C,which yields the lowest total number (N) of coarse and fine linearsearch operations used to determine the phase offsets of the trailingand leading edges of a data eye is given by the square root of 2M. Inthe example above in which M=512, expression (7) indicates a minimum Nwhen C=32. Inserting C=32 into expression (2) yields F1=F2=15. Similaranalyses may be performed for systems in which a coarse linear search isfollowed by a bounded binary search, and for systems in which phasecomparison operations are performed more than once per phase offset(e.g., performing the phase comparison operation multiple times tofilter erroneous lead-lag determinations).

For some values of M (256, for example), the expression (7) yields anon-integer value for C which, when rounded up or down to the nearestinteger, may lead to a value of N that is not a minimum. In such cases,neighboring values of C (i.e. C+1, C+2, . . . , C−1, C−2, . . . ) can bechecked to determine if the resulting number of searches (i.e. values ofN) are lower than with the calculated value of C. Alternatively, thecalculated, rounded value of C can be used since the corresponding valueof N will be close to the absolute minimum, if not the absolute minimum.

Timing Maintenance; Compensation for Drift

In one embodiment of the present invention, the fast phase-jumpingability of the locked loop circuit 200 of FIG. 3 (or locked loop circuit745 of FIG. 16) is employed to perform a periodic (and/or event-driven)timing maintenance operation, for example to compensate for a voltage-and/or temperature-induced phase error. Referring to FIG. 27, duringnormal operation of the locked loop circuit, a receive clock signal isused to sample an incoming data waveform in the center of eachsuccessive data eye 877 to provide maximum leading and trailing edgemargin (note that sampling instants offset from the center of the dataeye may be used in alternative embodiments, particularly where the datasetup and hold times of the receiver circuit are asymmetric). Due tochanges in voltage, temperature or other environmental or deviceparameters, the actual sampling instant, indicated by 878 may becomeskewed relative to the desired sampling instant, resulting in a loss oftiming margin.

Referring to FIGS. 27 and 28, in one embodiment, leading-edge andtrailing-edge phase offset values that correspond to leading andtrailing edge boundaries of the data eye 877 are recorded in a storageregisters 965 and 969 within a phase jumping locked loop circuitaccording to the invention (or elsewhere in the integrated circuit thatincludes the phase jumping locked loop circuit) and therefore may beselected for summation with the phase count value (i.e., generated bythe tracking loop 203 of FIG. 3 and the tracking loop 747 of FIG. 16) togenerate leading- and trailing-edge sampling clocks. In one embodiment,the leading-edge and trailing-edge phase offset values are generatedduring system initialization (e.g., by using the search operationsdescribed above in reference to FIGS. 23-26), and used to generate areceive phase offset value that is stored in register 967; the receivephase offset value being generated, for example, by averaging theleading-and trailing-edge offset values stored in registers 965 and 969.During normal operation, sample select signal 970 (SSEL) is set to anormal state to select, via select circuit 963, register 967 to source aphase offset value to be summed with the phase count value (PHASE CNT)in adder 971. The resulting phase control value 968 is then supplied tophase jumping mixer 751 which generates an offset clock signal(optionally to propagate through a clock tree) for timing the samplinginstant 878.

When a timing maintenance operation is to be performed, the sampleselect signal 970 is transitioned to a leading-edge-test state, andselects register 965 to source a leading edge phase offset value toadder 971. Consequently, the phase control value 968 is transitionedfrom the sampling clock phase control value to a leading-edge phasecontrol value. The phase jumping mixer 751 responds to the transition ofthe phase control value 968 by rapidly transitioning the phase of theoutput clock signal to the leading edge sampling instant shown at 955 ofFIG. 27. A data transfer test is then performed to determine whether anincoming data eye (or sequence of data eyes) is properly received whensampled at the leading fringe of the data eye 877 (i.e., at 955). If theleading-edge data transfer test is passed (i.e., proper data receptionconfirmed), the leading-edge phase offset value within register 965 isdecremented to establish the new leading-edge sampling instant shown at956. If the leading-edge data transfer test is failed, the leading-edgephase offset value within register 965 is incremented to establish thenew leading-edge sampling instant shown at 957.

After a pass/fail result is recorded for the leading-edge data transfertest, the sample select signal 970 is transitioned to atrailing-edge-test state, and selects register 969 to source a trailingedge phase offset value to adder 971. The adder 971 responds to the newphase offset value by transitioning the phase control value 968 from theleading-edge phase control value to a trailing-edge phase control value,and the phase jumping mixer 751 responds in turn by rapidlytransitioning the phase of the output clock signal to the trailing edgesampling instant shown at 959 of FIG. 27. A data transfer test is thenperformed to determine whether an incoming data eye (or sequence of dataeyes) is properly received when sampled at the trailing fringe of thedata eye 877 (i.e., at 959). If the trailing-edge data transfer test ispassed, the trailing-edge phase offset value within register 969 isincremented to establish the new leading-edge sampling instant shown at960. If the trailing-edge data transfer test is failed, thetrailing-edge phase offset value within register 969 is decremented toestablish the new leading-edge sampling instant shown at 961.

Still referring to FIG. 27, it can be seen that if the leading-edge datatransfer test passes and the trailing-edge data transfer test fails,then the phase of the data eye 877 has shifted in the direction of theleading edge (i.e., the phase of the data eye 877 has advanced relativeto the reference clock signal). Thus, the receive clock phase offsetvalue stored in register 967 is decremented in response to aleading-edge pass/trailing-edge fail condition, thereby keeping thesampling instant 878 substantially phase aligned with the center of thedata eye 877 (or phase aligned with a desired phase offset from thecenter of the data eye 877). Conversely, if the leading-edge datatransfer test fails and the trailing-edge data transfer test passes,then the data eye 877 has shifted in the direction of the trailing edge(i.e., the phase of the data eye has become increasingly delayedrelative to the reference clock signal), and the receive clock phaseoffset value stored in register 967 is incremented to keep the samplinginstant substantially phase aligned with the center of the data eye (orphase aligned with a desired phase offset from the center of the dataeye 877). By periodically repeating the phase adjustment operationsillustrated in FIG. 27, the sampling instant 878 is enabled to trackphase drift in the data eye (e.g., caused by changes in voltage andtemperature), thereby conserving system timing margin.

FIG. 29 is a flow diagram of a timing maintenance operation according toan embodiment of the invention. At 975 the leading-edge phase offsetvalue (i.e., the value stored in register 965 of FIG. 28) is selected togenerate a leading-edge-aligned clock signal. At 977, a leading-edgedata transfer test is performed. If the leading-edge data transfer testis determined to pass (979), then at 973, a Boolean variable, LE TEST isassigned a PASS value, and the leading edge phase offset value isdecremented. If the leading-edge data transfer test is determined not topass, then at 981, LE TEST is assigned a FAIL value, and the leadingedge phase offset value is incremented. At 985, the trailing-edge phaseoffset-value (i.e., the value stored in register 969 of FIG. 28) isselected to generate a trailing-edge-aligned clock signal. At 987, atrailing-edge data transfer test is performed. If the trailing-edge datatransfer test is determined not to pass (989), then at 991 the trailingedge phase offset value is decremented. The Boolean variable, LE TEST isinspected at 993 to determine whether the leading-edge phase offsetvalue has also been decremented (i.e., LE TEST =PASS) and, if so, thereceive clock phase offset value (i.e., the value stored in register 967of FIG. 28) is decremented to track the shift in the data eye. If theleading-edge phase offset value has not been decremented, the receiveclock phase offset value is not adjusted. If the trailing-edge datatransfer test is determined to pass, then the trailing-edge phase offsetvalue is incremented at 996, and the LE TEST variable is inspected at997 to determine whether the leading-edge phase offset value has alsobeen incremented (i.e., LE TEST=FAIL). If the leading-edge phase offsetvalue has been incremented, then the receive clock phase offset value isincremented at 999 to track the shift in the data eye. If theleading-edge phase offset value has not been incremented, then thereceive clock phase offset value is not adjusted. It should be notedthat the increment and decrement operations at 999 and 995,respectively, effectively maintain the receive clock phase offset valuemidway between the leading- and trailing-edge phase offset values. In analternative embodiment, the receive clock phase offset value may bere-calculated after each update to the leading- and/or trailing-edgephase offset value, for example, by dividing a sum of the leading- andtrailing-edge phase offset values by two (i.e., halving the sum of theoffset values) or by another predetermined number.

Per-Device Phase Offset; Source- and Destination-Based Phase Jumping

In one embodiment of the present invention, the fast phase-jumpingability of the locked loop circuit 200 of FIG. 3 (and locked loopcircuit 745 of FIG. 16) is employed to enable source- anddestination-based phase jumping. FIG. 30 illustrates a signaling device1000 which includes a storage circuit 1009 to store a number, N, oftransmit clock phase offset values, each transmit clock phase offsetvalue corresponding to a respective one of a plurality of remote deviceswithin a signaling system. In one embodiment, the storage circuit 1009includes a plurality of storage elements arranged in rows and columns.Access enable lines 1006 are coupled to respective rows of storageelements and bit lines (not shown in FIG. 30) are coupled to respectivecolumns of storage elements. When an address-selected one of the accessenable lines 1006 is activated, read or write access to thecorresponding row of storage cells is enabled, with the access toindividual storage elements of the row occurring via respective bitlines. By this arrangement, each of the rows of storage elements isenabled to store a respective phase offset value received via the bitlines in a write operation, and each of the rows of storage elements isenabled to output a previously stored phase offset value in a readoperation. When application logic 1003 receives (or generates) a requestto transmit data to one of the remote devices, the application logic1003 outputs a transmit identifier value 1002 which identifies theremote device intended to receive the transmission. The transmitidentifier is received within an address decoder 1005 which activatesone of the plurality of access-enable lines 1006 to enable acorresponding one of the transmit phase offset values (each being adigital value stored within a row of storage elements within the storagecircuit 1009) to be output to adder 1015 via bit lines 1010. Theselected transmit phase offset value is added to the phase count value1023 generated within the tracking loop 1001 to generate an updatedphase control word 1013. The phase jumping mixer 751 responds to theupdated phase control word 1013 by rapidly transitioning the phase of anoutput clock signal 1015 to the indicated transmit phase offset. Theoutput clock signal 1015 propagates through a clock tree circuit 1019(which may be omitted where significant clock signal fan out is notneeded) to generate a transmit clock signal 1018 having the desiredphase. By this arrangement, the locked loop circuit responds to each newtransmit identifier 1002 output by the application logic (andcorresponding transmit phase offset value output from the storagecircuit 1009) by rapidly transitioning the phase of the transmit clocksignal 1018 to the transmit phase offset recorded for the correspondingremote device.

Still referring to FIG. 30, storage circuit 1011 is provided to store anumber, N, of receive clock phase offset values, each transmit clockphase offset value corresponding to a respective one of the plurality ofremote devices within the signaling system. When application logic 1003receives (or generates) a request to receive data from one of the remotedevices, the application logic 1003 outputs a receive identifier value1004 which identifies the remote device. The receive identifier isreceived within an address decoder 1007 which activates one of aplurality of access-enable lines 1008 to enable a corresponding one ofthe receive phase offset values (each being a digital value storedwithin a row of storage elements within the storage circuit 1011) to beoutput to adder 1017 via bit lines 1012. The selected transmit phaseoffset value is added to the phase count value 1024 to generate anupdated phase control word 1014. The phase jumping mixer 751 _(RX)responds to the updated phase control word 1014 by rapidly transitioningthe phase of an output clock signal 1016 to the indicated receive phaseoffset. The output clock signal 1016 propagates through a clock treecircuit 1021 (which may be omitted where significant clock signal fanout is not needed) to generate a receive clock signal 1020 having thedesired phase. By this arrangement, the locked loop circuit responds toeach new receive identifier 1004 output by the application logic (andcorresponding receive phase offset value output from the storage circuit1011) by rapidly transitioning the phase of the receive clock signal1020 to the receive phase offset recorded for the corresponding remotedevice. Note that the application logic 1003 may generate the request toreceive data from a remote device in response to a previous transmissionto the remote device. For example, in a memory system, the signalingdevice 1000 may be a memory controller that transmits a memory readcommand (or memory read request) to a remote memory device, the readcommand evoking a responsive, deterministic transmission by the memorydevice that is received by one or more receive circuits within thesignaling device 1000 under control of the receive clock signal 1020.

Still referring to FIG. 30, the storage circuits 1011 and 1009 andcorresponding address decoders 1005 and 1007 may be replaced by aunified storage circuit and corresponding unified address decoder in analternative embodiment. In such an embodiment, the application logic1003 may output a device identifier to indicate which device is to bethe source or destination of a transmission, with a most significant bit(or least significant bit of the device identifier being used to selectbetween transmit and receive phase offsets. Also, while separate adders(1015, 1017), phase jumping mixers (751 _(TX), 751 _(RX)) and clock treecircuits 1019 and 1021 are depicted in FIG. 30, a single adder (orsumming circuit 767 of FIG. 16), phase jumping mixer and-clock treecircuit may be alternatively be used in the arrangement described inreference to FIGS. 16-18 to alternatively generate a transmit clocksignal (having a phase offset according to the selected one of Ntransmit phase offset values) and a receive clock signal (having a phaseoffset according to the selected one of N receive phase offset values)on a shared clock line.

Locked Loop Circuit with Clock Hold Function

FIG. 31 illustrates a phase-jumping locked loop circuit 1101 thatgenerates a clock signal 1130 for clocking a synchronous logic circuit1103. The locked loop circuit 1101 includes a tracking loop 1105,reference loop 1107, and offset clock generator 1109. The reference loop1107 operates as described above in reference to FIGS. 3 and 10 tooutput a plurality of phase vectors 1110 (PV) to phase mixing circuitswithin the tracking loop 1105 and offset clock generator 1109. Thetracking loop 1105 operates as described above in reference to FIGS. 3and 16 to adjust a phase count 1112 (PCNT) as necessary to produce afeedback clock signal 1108 (FCLK) that is phase-aligned with a referenceclock signal 202. The phase count value 1112 represents a phase offsetbetween the feedback clock signal and a reference phase vector (i.e.,one of the phase vectors 1110 designated to have, for example, a zerodegree phase angle), and is supplied to the offset clock generator 1109along with the feedback clock signal 1108.

The offset clock generator 1109 includes an offset selector 1121, adder1131, phase jumping mixer 1123, clock hold circuit 1125, and clock treecircuit 1127. The offset selector 1121 selects between offset values1114 and 1116 (OFST1 and OFST2, respectively) according to an offsetselect signal 1102 (OFF_SEL). The selected offset value 1118 is summedwith the phase count value 1112 in adder 1131 to produce a phase controlvalue 1120. In an alternative embodiment more than two offset values maybe input to the offset selector 1121, and the offset values or a subsetthereof may be maintained within the locked loop circuit 1101 ratherthan being provided by external logic. Also, the phase count value 1112may alternatively be summed with each of the offset values in separateadder circuits, with the summed values being input to the offsetselector 1121 (see summing circuit 750 of FIG. 16, for example).Further, while the offset selector 1121 is depicted as a multiplexer inFIG. 31, any circuit capable of selecting one of a plurality of offsetvalues or phase control values (e.g., an address decoder in associationwith a memory array, register file or other storage), may be used inalternative embodiments.

The phase jumping mixer 1123 generates a mix clock signal 1122 (MCLK) byinterpolating between a selected pair of the phase vectors 1110 inaccordance with the phase control value 1120. The mix clock signal 1122is provided to the clock hold circuit 1125 which, in response, outputs ahold clock signal 1124 (HCLK) to the clock tree circuit 1127. The holdclock signal 1124 propagates through the clock tree circuit 1127 toproduce multiple instances of an offset clock signal, at least one ofwhich is the clock signal 1130 provided to the synchronous logic circuit1103. The synchronous logic circuit includes one or more logic circuitswhich respond to transitions in the clock signal 1130 (e.g., flip-flopsand/or other edge-triggered logic circuits). Note that the clock treecircuit 1127 may be omitted in embodiments in which the fan-out of thehold clock signal 1124 is limited. Also, a delay circuit which exhibitssubstantially the same propagation delay as the clock hold circuit 1125may be included within the tracking loop 1105 such that, in the case ofa zero-valued offset 1118, clock signal 1130 is substantially phasealigned to with the feedback clock signal 1108.

FIG. 32 illustrates an exemplary relationship between the offset selectsignal 1102, mix clock signal 1122 and hold clock signal 1124. Alsoshown are two clock signals, OFST1 CLK (1142) and OFST2 CLK (1144),which correspond to mix clock signals 1122 that will be generated by thephase jumping mixer 1123 for corresponding selections of the offsetcontrol values 1114 and 1116 (OFST1 and OFST2, respectively). To beclear, clock signals OFST1 CLK and OFST2 CLK are not separatelygenerated within the locked loop circuit 1101, but rather representinstances of the mix clock signal 1122 that correspond to OFST1 andOFST2, respectively. Thus, when the offset select signal 1102 is low,OFST1 is selected as the offset control value 1118, and the mix clocksignal 1122 has a phase according to the OFST1 CLK 1142. When the offsetselect signal 1102 goes high at 1152, a phase jump is initiated withinthe phase jumping mixer to transition the phase of the mix clock signal1122 from the phase of OFST1 CLK 1142 to the phase of OFST2 CLK 1144.Because of the phase difference between OFST1 CLK and OFST2 CLK, thephase jump produces a short-duration pulse 1154 within the mix clocksignal 1122, referred to herein as a runt pulse. Depending on thestarting time and duration of the phase jump operation, the runt pulse1154 may be wider or narrower than shown in FIG. 32, and may be alow-level runt pulse rather than a high-level runt pulse.

Clock signals exhibiting occasional runt pulses may be tolerated in somesystems, (e.g., where the clock signal is used solely to clockinput/output circuits), but tend to produce undesirable meta-stablestates and/or race conditions in synchronous logic circuits due to theinability to guarantee signal setup and hold times and due to theuncertain transition time of state variables (e.g., flip-flop outputs).In the locked loop circuit 1101 of FIG. 31, the clock hold circuitsuppresses runt clock pulses to avoid meta-stability and raceconditions.

Still referring to FIGS. 31 and 32, the clock hold circuit 1125 iscoupled to receive the offset select signal 1102 and, upon detecting atransition in the offset select signal 1102, latches the state of thehold clock signal 1124 over a clock hold interval 1156 that is longenough to avoid generation of a low-level or high-level runt pulse. Inone embodiment, the mix clock signal 1122 may have an arbitrary phaserelative to the transition time of the offset select signal, andtherefore the hold clock signal 1124 may be in transition at the startof the clock hold interval 1156. To prevent latching or otherwisecapturing an indeterminate state of the hold clock signal 1124, theclock hold circuit includes circuitry to predict whether the leadingedge of the clock hold interval 1156 will coincide with a transition ofthe hold clock signal 1124 and, if so, to delay the start of the clockhold interval to a later time, thereby producing a delayed clock holdinterval 1158. In this way, a determinate state of the hold clock signal1124 will be latched by the clock hold circuit 1125, regardless of whenthe offset select signal 1102 transitions.

FIG. 33 illustrates the clock hold circuit 1125 of FIG. 31 according toan embodiment of the invention. The clock hold circuit 1125 includes alatch 1175, hold signal generator 1177, synchronizing logic 1181, andkeepout signal generator 1179. The latch 1175 receives the mix clocksignal 1122 from a phase jumping mixer and, so long as a qualified boldsignal 1194 is deasserted (i.e., at a latch-enable input, LE), passesthe mix clock signal 1122 to the latch output (Q) as the hold clocksignal 1124. When the qualified hold signal 1194 is asserted, the latch1175 maintains (i.e., latches) the hold clock signal 1124 at its mostrecently output state, even as the mix clock signal 1122 changes stateat the input of latch 1175.

The hold signal generator 1177 includes a hold control circuit 1183,delay element 1187 (D₁), exclusive-OR gate 1185, delay element 1189 (D₂)and multiplexer 1191. In one embodiment, the hold control circuit 1183is a finite state machine that outputs a hold signal 1190 as a statevariable, and that transitions between states according to therespective states of a jump signal 1202, and a clock-XOR signal 1186.The jump signal 1202 is asserted by the synchronizing logic 1181 inresponse to a transition in the offset select signal 1102, and thereforeindicates that a phase jump in the mix clock signal 1122 is beinginitiated. The clock-XOR signal 1186 is high whenever the hold clocksignal 1124 and a delayed instance 1188 of the mix clock signal 1122(i.e., delayed by delay element 1187), have different states. In oneembodiment, the delay element 1187 is formed by an inverter chain thatmatches an inverter chain in a non-latching input-to-output path withinthe latch 1175. Consequently, when the qualified hold signal 1194 isdeasserted, the delayed mix clock signal 1188 is phase aligned with thehold clock signal 1124, and the clock-XOR signal 1186 is low. Bycontrast, when the qualified hold signal 1194 is asserted, the clock-XORsignal goes high when the delayed mix clock signal 1188 transitions to astate different from the latched state of the hold-clock signal. Thatis, the clock-XOR signal goes high at the start of the first high- orlow-level pulse following assertion of the qualified hold signal 1194.

FIG. 34 is an exemplary state diagram of the hold control circuit 1183of FIG. 33. Referring to FIGS. 33 and 34, the hold control circuit 1183is initialized to a first state 1251 and remains in state 1251 until thejump signal is asserted. State 1251 is a non-hold state, meaning thatthe hold signal 1190 is deasserted and therefore that the qualified holdsignal 1194 is deasserted and the hold clock signal 1124 tracks the mixclock signal. When the jump signal 1202 is asserted, the hold controlcircuit 1183 transitions to a second state 1253. In state 1253, the holdsignal 1190 is asserted, producing a corresponding assertion ofqualified hold signal 1194 to latch the state of the hold clock signal1124. When the hold signal 1190 is initially asserted, the state of thehold clock signal and the delayed mix clock signal 1188 are the same sothat the clock-XOR signal 1186 is low. At the first transition of thedelayed mix clock signal 1188 following assertion of qualified holdsignal 1194, the states of the delayed mix clock signal 1188 and thelatched hold clock signal 1124 will diverge, thereby causing theclock-XOR signal 1186 to go high. The hold control circuit 1183 respondsto the high-going clock-XOR signal 1186 by transitioning to a thirdstate 1255 in which the hold signal 1190 (and therefore the qualifiedhold signal 1194) remains asserted. Referring briefly to FIG. 32, it canbe seen that the first transition of the mix clock signal 1122 followingthe start of a clock hold interval (1156 or 1158) is a leading edge(rising or falling) of a potentially short-duration pulse (i.e., apotential runt pulse). The hold control circuit 1183 remains in state1255 while the both the clock-XOR signal 1186 and the jump signal 1202are high. After the second transition of the delayed mix clock signal1188 (i.e., a trailing edge of the potential runt pulse), the delayedmix clock signal 1188 again matches the state of the latched hold clocksignal 1124 so that the clock-XOR signal 1186 goes low. The hold controlcircuit 1183 transitions to a fourth state 1257 in response to thelow-going clock XOR signal. In state 1257, the hold signal 1190 isdeasserted, resulting in a corresponding deassertion of the qualifiedhold signal 1194 and restoration of the latch 1175 to a non-latchedcondition. Thus, after the potential runt pulse within the mix clocksignal 1122 has passed, the hold clock signal 1124 is enabled tocontinue tracking the mix clock signal 1122. When the jump signal 1202is deasserted, the hold control circuit 1186 returns to the initialstate 1251. In one embodiment, illustrated in FIG. 34, the hold controlcircuit 1186 is further adapted to transition to state 1251 from anyother of the states (1253, 1255 or 1257) in response to a low going jumpsignal 1202.

As briefly discussed above, if the qualified hold signal 1194 isasserted coincidentally with a transition of the hold clock signal 1124(or transition of the mix clock signal 1122 or an intermediary clocksignal generated within the latch 1175), a metastable hold clock signal1124 may be output by latch 1175 (i.e., the voltage level of hold clocksignal 1124 may fall within an invalid range between two valid outputvoltage levels). In addition to the potential for producing undesiredresults in the synchronous logic circuit 1103 of FIG. 31, a metastablehold clock signal 1124 will potentially produce a metastable XOR-clocksignal 1186 and therefore disrupt the operation of the hold controlcircuit 1183 and the clock hold circuit 1125 generally. The keepoutcircuit 1179 of FIG. 33 is provided to prevent such undesired results.

Referring to FIG. 33, the keepout circuit 1179 includes delay elements1201 (D₃) and 1203 (D₄), exclusive-OR gates 1205 and 1207, AND gates1211 and 1213 and set/reset (S-R) flip-flop 1215. A jump-test signal1204 (JTST) is supplied to a first input of exclusive-OR gate 1207 andto an input of delay element 1203. The output of delay element 1203 issupplied to a second input of exclusive-OR gate 1207 so that eachtransition of the jump test signal 1204 causes exclusive-OR gate 1207 tooutput a pulse 1208. The duration of pulse 1208 corresponds to thepropagation delay through delay element 1203 and defines a time intervalreferred to herein as a jump window (JWIN). The mix clock signal 1122 issupplied to a first input of exclusive-OR gate 1205 and to an input ofdelay element 1201. The output of delay element is supplied to a secondinput of exclusive-OR gate 1205 so that each transition of the mix clocksignal 1122 causes exclusive-OR gate 1205 to output a pulse 1206. Theduration of pulse 1206 corresponds to the propagation delay throughdelay element 1201 and defines a time interval referred to herein as aclock window (CWIN). In one embodiment, delay element 1201 produces asubstantially longer delay than delay element 1203 (e.g., by including alonger chain of inverters or other delay circuits) so that the clockwindow is substantially wider than the jump window.

FIG. 35 illustrates exemplary timing relationships between the clockwindow and the jump window defined respectively by signals 1206 and 1208of FIG. 33. Because the leading edge of the clock window is generated inresponse to a transition of the mix clock signal 1122, the clock windowrepresents a time interval during which assertion of the jump signal1202 may result in coincident transitions in the hold clock signal 1124and the qualified hold signal 1194. In one embodiment, the jump testsignal 1204 is a periodic signal that is phase aligned with the jumpsignal 1202 so that the jump window corresponds to a potential assertiontime of the jump signal 1202 (i.e., if the offset select signal istransitioned). Thus, as indicated in FIG. 35, if the jump window fallswithin the clock window, a keepout signal 1216 is asserted. Referring toFIG. 33, the output of S-R flip-flop 1215 constitutes the keepbutsignal, 1216. The S-R flip flop 1215 is initially in a reset state inwhich the keep out signal 1216 is deasserted. When signals 1206 and 1208are both high (i.e., the jump window falls at least partially within theclock window), the output of AND gate 1211 goes high to set the S-Rflip-flop and thereby assert the keepout signal 1216. Multiplexer 1191within the hold signal generator 1177 responds to the asserted keepoutsignal 1216 by selecting a delayed hold signal 1192 (i.e., generated bypropagation of hold signal 1190 through delay element 1189 (D₂)) to beoutput as the qualified hold signal 1194 to the latch 1175. If the jumpwindow falls outside the clock window, then signal 1208 will be highwhile signal 1206 is low, causing AND gate 1213 to reset the S-R flipflop and thereby deassert the keepout signal 1216. The multiplexer 1191responds to the deasserted keepout signal 1216 by selecting the holdsignal 1190 to be output as the qualified hold signal 1194. Thus, thedelayed hold signal 1192 is output as the qualified hold signal 1194when the relative transition times of the jump test signal 1204 and themix clock signal 1122 indicate a likelihood that an assertion of thehold signal 1190 will coincide with a transition in the mix clock signal1122 (and therefore with a transition in the hold clock signal 1124).Conversely, the hold signal 1190 is output as the qualified hold signal1194 when the relative transition times of the jump test signal and themix clock signal indicate that an assertion of the hold signal 1190 willnot coincide with a transition in the mix clock signal 1122.

FIG. 36 is an exemplary state diagram of the keepout circuit 1179 ofFIG. 33. Referring to both FIGS. 36 and 33, the keepout circuit 1179 isinitialized to a first state 1275 in which the keepout signal isdeasserted. The keepout circuit 1179 remains in state 1275 so long asthe jump window and clock window do not overlap (i.e., so long as theBoolean expression /JWIN OR /CWIN remains true). If the jump window andclock window overlap (i.e., signals 1206 and 1208 are both high), theS-R flip-flop 1215 is set, transitioning the keepout circuit 1179 tostate 1277, in which the keepout signal 1216 is asserted. The keepoutcircuit 1179 remains in state 1277 so long as the jump window does notfall outside the clock window (i.e., so long as the Boolean expression/JWIN OR CWIN remains true). If the jump window falls outside the clockwindow (i.e., signal 1206 is low while signal 1208 is high), the S-Rflip-flop 1215 is reset, returning the keepout circuit to state 1275 andtherefore deasserting the keepout signal 1216.

FIG. 37 illustrates an exemplary embodiment of the synchronizing logic1181 of FIG. 33. The synchronizing logic 1181 includes flip-flops 1281,1283 and 1285, and an exclusive-OR gate 1289. Each of the flip-flops(1281, 1283, 1285) is clocked by the feedback clock signal 1108 (i.e.,generated within the tracking loop 1105 of FIG. 31). The offset selectsignal 1102 is supplied to a data input of flip-flop 1281 and to a firstinput of exclusive-OR gate 1289. The output of flip-flop 1281 issupplied to a second input of the exclusive-OR gate 1289 so that, whenthe offset select signal 1102 changes state, exclusive-OR gate 1289asserts a jump detect signal 1290 until the next rising edge of thefeedback clock signal 1108. The output of the exclusive-OR gate 1289 iscoupled to a data input of flip-flop 1283 so that the asserted jumpdetect signal 1290 is registered within flip-flop 1283 in response tothe rising edge of the feedback clock signal that succeeds thetransition in the offset select signal 1102. The jump signal 1202 isoutput via an inverting output of the flip-flop 1283 and thereforeconstitutes a synchronous indication of the offset select signaltransition. Note that the offset select signal 1102 may be amultiple-bit signal (e.g., used to select between more than two offsetcontrol values or phase control values) in which a transition within anybit of the offset select signal 1102 results in assertion of the jumpsignal 1202. Still referring to FIG. 37, the input and inverting outputof flip-flop 1285 are coupled to one another to generate the jump testsignal 1204. Thus, in the embodiment of FIG. 37, the jump test signal1204 is a periodic signal that transitions in response to each edge ofthe feedback clock signal 1108 and that is substantially phase alignedwith a transition in the jump signal 1202. In alternative embodiments,clock signals other than the feedback clock signal 1108 may be used togenerate the jump and jump test signals (1202 and 1204), and fallingrather than rising edges of the feedback clock signal 1108 (or otherclock signal) may be used to trigger state changes within the flip-flops1281, 1283 and 1285.

It should be noted that the exemplary phase jumping applicationsdescribed above, though described in terms of phase jumping locked loopcircuits that include the phase jumping mixer embodiments described inreference to FIGS. 4-14, may alternatively be implemented by locked loopcircuits that include other types of mixing circuits. In general, anycircuit of producing a relatively rapid phase transition in an outputclock signal may be used in the above-described applications in place ofthe phase jumping mixer embodiments described in reference to FIGS.4-14.

The section headings provided in this detailed description are forconvenience of reference only, and in no way define, limit, construe ordescribe the scope or extent of such sections. Also, while the inventionhas been described with reference to specific exemplary embodimentsthereof, it will be evident that various modifications and changes maybe made thereto without departing from the broader spirit and scope ofthe invention. Accordingly, the specification and drawings are to beregarded in an illustrative rather than a restrictive sense.

1-25. (Canceled)
 26. A method comprising: outputting a phase adjustsignal in response to a first clock signal and a second clock signal;obtaining a phase count value in response to the phase adjust signal;summing a phase jump value and the phase count value in response to thephase adjust signal to obtain an updated phase count value; and biasinga plurality of transistors in a phase mixer in response to the updatedphase count value.
 27. The method of claim 26, wherein the first clocksignal is a feedback signal and the second clock signal is a referenceclock signal.
 28. The method of claim 27, wherein the phase jump valuerepresents a portion of a cycle of the referenced clock signal.
 29. Themethod of claim 27, wherein the phase jump value is a negative value.30. The method of claim 27, further comprising: determining whether thefeedback signal leads the reference clock signal; and determiningwhether the feedback signal lags the reference clock signal.
 31. Themethod of claim 27, further comprising: reducing the phase jump value;and repeating the outputting, obtaining and summing.
 32. The method ofclaim 26, wherein the obtaining is performed by a phase counter and theoutputting is performed at by a phase detector.
 33. The method of claim32, further comprising: disabling the counter.
 34. The method of claim33, further comprising: enabling the counter.
 35. A method comprising:disabling a phase counter to adjust a phase count value; initializing aphase jump value; comparing a reference clock signal with a feedbackclock signal; summing the phase jump value and the phase count value toobtain an updated phase count value in response to comparing thereference clock signal to the feedback signal; loading the updated phasecount value into the phase counter; comparing the phase jump value to athreshold value; and reducing the phase jump value in response tocomparing the phase jump value to the threshold value.
 36. The method ofclaim 35, further comprising: biasing a plurality of transistors in aphase mixer in response to the updated phase count value.
 37. The methodof claim 35, further comprising: repeating the comparing the referenceclock signal with the feedback clock signal; repeating the summing thephase jump value and the phase count value; repeating the loading theupdated phase count value; repeating the comparing the phase jump valueto the threshold value; and repeating the reducing the phase jump value.38. The method of claim 35, further comprising: enabling the phasecounter in response to the comparing the phase jump value to thethreshold value.
 39. The method of claim 35, wherein the phase jumpvalue is initialized to represent one fourth of a cycle of the referenceclock signal.
 40. The method of claim 35, wherein the phase jump valueis a negative value.
 41. An integrated circuit device comprising: aselect circuit to provide a first offset value; a summing circuit to sumthe first offset value and a phase count value, wherein the phase countvalue indicates a phase difference between a reference clock signal anda clock signal in a plurality of clock signals; and a phase mixer tocombine the first plurality of clock signals with the sum of the firstoffset value and the phase count value, wherein the phase mixerincludes: a plurality of differential amplifiers including a first,second, third and fourth differential amplifier, a biasing circuit,coupled to the plurality differential amplifiers, including a first,second, third and fourth biasing transistor, wherein the first andsecond biasing transistors are coupled in series with the first andsecond differential amplifiers, and the third and fourth biasingtransistors are coupled in series to the third and fourth differentialamplifiers.
 42. The integrated circuit of claim 41, wherein theplurality of differential amplifiers has a plurality of inputs andoutputs, wherein the plurality of inputs receives the plurality of clocksignals and the plurality of outputs are coupled to a first and secondoutput signal line.
 43. The integrated circuit of claim 42, where afirst resistive element is coupled to a first reference voltage and thefirst output signal line.
 44. The integrated circuit of claim 43,wherein the first resistive element has a variable resistance value inresponse to a first control signal.
 45. The integrated circuit of claim43, wherein a second resistive element is coupled to the first referencevoltage and the second output signal line.
 46. The integrated circuit ofclaim 45, wherein the second resistive element has a variable resistancevalue in response to the first control signal.
 47. The integratedcircuit of claim 41, wherein a first differential amplifier in theplurality of differential amplifiers includes a first transistor havinga drain coupled to a first output signal line and a second transistorhaving a drain coupled to a second output signal line.
 48. Theintegrated circuit of claim 47, wherein the first transistor has a firstsource and the second transistor has a second source coupled to thefirst source.
 49. The integrated circuit of claim 48, wherein the firstand second sources are coupled to a switch element that is coupled tothe biasing circuit.
 50. The integrated circuit of claim 41, wherein thefirst biasing transistor is coupled in series to a fifth biasingtransistor and the second biasing transistor is coupled in series to asixth biasing transistor.
 51. An integrated circuit, comprising: a phasemixer having a plurality of differential amplifiers to combine a pair ofclock signals; means for current biasing a first differential amplifierin the plurality of differential amplifiers using a first plurality ofbiasing transistors and a second differential amplifier in the pluralityof differential amplifiers using a second plurality of biasingtransistors.
 52. The integrated circuit of claim 51, further comprising:a switch element coupled to the first differential amplifier and thefirst plurality of biasing transistors.